李健榮助理教授

Department of Electronic EngineeringNational Taipei University of Technology

Multiband RF Transceiver System Chapter 6 Multi-mode and

Multi-band Transceivers

Outline

• System Level Considerations

• Wideband LO Generation

• Building Blocks of TX and RX

• Isolating Techniques on ICs

• RF Power Amplifier

• Two multi-mode examples of the transceiver areintroduced in this chapter.

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Software-Defined Front-ends

• The ultimate dreamof every SDR front-end is to deliver an RFtransceiver that can be reconfigured into every operating mode.

Modes for cellular (2G–2.5G–3G and further), WLAN (802.11a/b/g/n), WPAN(Bluetooth, Zigbee, etc.), broadcasting (DAB, DVB, DMB, etc.), and positioning(GPS, Galileo) functionalities. Obviously, each of them has different centerfrequency, channel bandwidth, noise levels, interference requirements, transmitspectral mask, and so on.

• As a consequence, the performances of all building blocks inthe transceiver mustbe reconfigurable over an extremelywide range.

Linearity, filtering, noise, bandwidth, and so on, can be traded for powerconsumption.

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System Level Considerations

• A first choice to be made is the radio architecture to be used.

Heterodyne, homodyne, low-IF, and other architectures, which one to choose?

• In view of SDR, this question perhaps becomes a little easierto answer.

� When the characteristics of all possible standards are taken into account,not a single IF can be found that suits them all.

� Having multiple IFs and the associated (external) filtering stages increasesthe hardware cost of the SDR, which cannot be tolerated.

� Thus, thedirect-conversion architectures are the right choice for the job.

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Vision of the SDR Transceiver

• The transceiver core implemented in CMOS includes a fullyreconfigurable direct conversion RX, TX, and two synthesizers(for FDD).

� The functions that cannot beimplemented in CMOS areincluded on the package substrate.

� These are related primarily to theinterface between the active coreand the antenna. They mustprovide high-Q bandpass filteringor even duplexing, impedance-matching circuits, and poweramplification.

MEMS switches

Tunable matching

Tunable filtering

Power amplifier

DMQ

VCODistr.

DMQ

NoC controller

Frac-NPLL

Frac-NPLL

MCM substrateCMOS IC

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Hard Works

• Determine performance specifications for each block.

• The total budget for gain, noise, linearity, and so on, must bedivided over all blocks, ensuring that all possible test cases arecovered for every standard.

• Having very flexible building blocks helps a great deal, butmaking a smart systemanalysis is crucial to obtaining anoptimal solution.

• Gain ranges and signal filtering must be set such that the signallevels are an optimal trade-off between noise and distortion.

• With the built-in flexibility, a software-defined radio canachieve state-of-the-art performance very close to that ofdedicated single-mode solutions.

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Wideband LO Synthesis

• Example:

To generate all required LOsignals in the range 0.1 to 6 GHz,several frequency generation techniques have been proposedto relax the tuning range specifications of a voltage-controlledoscillator (VCO).

They use division, mixing, multiplication, or a combination of these. However,to make these systems efficient in terms of phase noise and power consumption,the VCO tuning range still has to be maximized.

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Frequency Tuning Capacitor

• Frequency tuning ofLC VCOs is commonly done by changingthe capacitance value of the varactors and/or an array ofswitched capacitors in the tank.

Switched or controlled inductor designs remain difficult to cover the desired widebandcontinuously and to limit the deterioration of the phase noise performance caused by theinsertion of these switches.

• Instead of using a single large varactor to tune the frequency, amixed discrete/continuous tuning scheme is usually chosen.

A small varactor is used for fine continuous tuning, and larger steps are realized bydigitally switching capacitors in and out of the resonant tank.

This has two advantages: The VCO gain is lower, allowing easier phase-locked loop(PLL) design, and digitally switched varactors have a higher ratio between thecapacitance in the on-state (Con) and the capacitance in the off-state (Coff ). A higherCon/Coff ratio allows a larger VCO frequency tuning range.

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Tank Loss Variations

• In the target frequency range (< 5 GHz), the losses in theoscillator tank are usually dominated by the inductor.

This simplification is, of course, not completely valid, since extra lossesdue to theskin effect, for examples will increase the resistance at higher frequencies.

• The negative resistance needed to compensate for the inductorlosses is given byGm = RS(ωC)2.

• If we want the oscillation frequency to change by a factor of 2:

� Total capacitance of the resonant tank has to be changed by a factor of 4

� Required negative resistance must also change by a factor of 4.The transconductance required for the active core is four times higher at thelower end of the frequency tuning range than at the higher end.

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Wideband VCO Architecture

� Scale the core biasing current.

� Change the transistors sizes.

� Keep parasitics at a minimum (phase noise and thetuning range achievable).

� Switches to turn transistors on or off.

� Switches has to avoid degrading the oscillatorphase noise as well as to ensure parasiticcapacitances are small.

� For lower frequencies, more and more core unitsare gradually activated, and the total bias currentincreases to keep the oscillation amplitude steadyand the parasitic capacitance increases, helping the“normal” varactors in their goal to increase thetotal tank capacitance.

M1 M2

SW1 SW1M3 M4

Dunit

CkvcoVtune

0

1

1

0

Dtune

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Frequency Tuning Sensitivity Variations

• Variation in VCOsensitivity for wide-tuning-range VCOs:

A change in the control voltage Vtune results in a change C in the analog varactorcapacitance Cvar . This causes a change in frequency f:

• The VCOwith a frequency ratio of 2, tank capacitance has tochange by a factor of 4, the VCOfrequency sensitivity willthen change by a factor 4√4 = 8.

Such a large change in VCO gain presents serious problems for the design of thePLL in which it will be incorporated. It prevents keeping the PLL bandwidthconstant and hence endangers the loop stability and an optimal phase noiseperformance.

1 1

2 4

ff

CLC C LCπ π∆ −= ⇒ =∆

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0.1 to 6-GHz Quadrature Generation

• The divide/multiply and quadrature (DMQ) contains severaldivide-by-2 blocks. They generateI and Q phases down to adivision factor of 32.

DIV2DIV2DIV2DIV2DIV2

PPF

PPF

DIV2

BUF

4G

5G:3G

2G

4G:2

G

4G4G 1G 0.5

G

250

M

1.5G

125

M

• The DMQ further employs aquadrature-phase single-sideband (SSB) mixer to generate3GHz, 5GHz.

• The SDR’s LOfrequency canbe selected by a multiplexerintegrated in the DMQ.

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Receiver Building Blocks

• A key aspect for the receiver RF part is its interferencerobustness. The blocking requirements for simultaneous multi-mode operation imply the need for tunable narrowband circuitsat the antenna interface.

Either this function can be provided by a multi-band filtering block, in which casethe receiver’s input can be a wideband LNA, or part of this burden can be taken upin the LNA design.

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MEMS-Enabled Dual-Band LNA (I)

• Using MEMSs switches to build a low-loss reconfigurableantenna filter section on a thin-filmsubstrate.

Packaged MEMS switch :connect the LNA to eitherits 1.8 GHz or 5 GHzmatching circuit andantenna filter. The loss ofthe switch is only 0.2 dB.

Cx reduces the gate inductance for the input matching.

At 1.8-GHz, a simplematching made up of oneor two passive componentscan fulfill the matchingrequirement.

The bondpad is modeledby a 65-fF cap in serieswith a 50-Ohm resistance.Each bondwire is modeledby a 1.3-nH inductance.

InputStage

Singleto

Diff.Conv.

A

BA

B

GainCtrl

5-6GHz

1.8GHz

Cx

5-6 GHz

MEMSSPDT

1.8 GHz

C bp

Low-band

50 -TL MatchingNetwork

Lbond

Switchable MatchingNetwork On board On chip

High-band

Freq. [GHz]0 1 2 3 4 5 6 7 8

0-2-4-6-8

-10-12-14-16-18-20

S11

[d

B]

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MEMS-Enabled Dual-Band LNA (II)

Internally, the LNAhas two separateoutputs to cover therequired frequencyrange.

A resistively loaded output issmall in area and widebandwidth but can only provideenough gain at frequencies upto 2.5 GHz

This output is for the 5 to 6-GHzband with an LC-tuned load. Aresistor in parallel with this inductorlowers its Q to cover the 1-GHzbandwidth.

Gain switching: When the third CG-transistor is activated, which bypasses acertain fraction of the signal current tothe power supply so as to reduce the gain.

The overall gain is24 dB. S11 inputmatching betterthan −10 dB isachieved in bothbands. Thesimulated LNA NFis around 2 dB,while the IIP3value is −5 dBm inthe low band and 3dBm in the highband.

InputStage

Singleto

Diff.Conv.

A

BA

B

GainCtrl

5-6GHz

1.8GHz

Cx

5-6 GHz

MEMSSPDT

1.8 GHz

C bp

Low-band

50 -TL MatchingNetwork

Lbond

Switchable MatchingNetwork On board On chip

High-band

Freq. [GHz]0 1 2 3 4 5 6 7 8

0-2-4-6-8

-10-12-14-16-18-20

S11

[d

B]

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Wideband LNAs (I)

• Rely on the passives in the antenna interface for RFinterference and blocking filtering. This makes the realizationof the concept easier, as commercially available (multi-band)filtering blocks can be used in the implementation.

• Wideband LNAmust nowbe used that cover an RF frequencyrange as large as possible for optimal flexibility, but must stillachieve state-of-the-art performance with respect tonarrowband LNAs.

• Covering the full 100 MHz to 6 GHz frequency range ischallenging since achieving a lowNF at hundreds of MHzrequires large transistors with low1/ f noiseOn-chip LC-matched common-source (CS) LNAs typically cover a bandwidth from3 to 10 GHz. Extending the bandwidth down to 100 MHz would requireprohibitively large inductors and thus chip area.

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Wideband LNAs (II)

• Two LNAs are combined to cover the entire frequency range:

An inductor-less feedback LNA with a small form factor covers frequencies from100 MHz to 2.5 GHz, and a CS LC-matched LNA covers frequencies from 2.5 to 6GHz. Only one LNA is powered at a time, to save power and provide filtering overhalf of the bandwidth.

Resistive Feedback LNA

OUT

IN

C

Ba

ndg

ap

OUT

IN

VC

Ba

nd

gap

LC-Matched LNA

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Resistive Feedback LNA – 0.1 ~ 2.5 GHz

A digitally controlled bank of feedbackresistors allows us to switch from high-to intermediate- and low-gain modes.

The biasing is done witha 3-bit programmablecurrent source. Thisallows us to vary thegain in small stepsaround the different gainmodes and to decreasingthe power by half whenswitching from high- tolow-gain mode.

At a maximum gain of 22 dB, typical simulation results achieve an NF of 2 dBand an IIP3 of −10 dBm at a power consumption of 12 mW. At reduced gain(10 dB), the linearity improves to +3 dBm while the power consumptiondecreases to 8 mW.

OUT

IN

C

Ba

ndg

ap

It employs resistive feedback for wideband matching andnoise canceling for low NF over a wide band. It in generalhas lower gain and a higher noise figure than these ofinductively matched narrowband designs, but it offers largesavings in area.

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LC-Matched LNA – 2.5 ~ 6 GHz

Broadband input-matching is achieved by the inductively degenerated CS-stage into anLC bandpassfilter . Input matching from 6 GHz down to 2.4 GHz can be done with inductive elements of reasonablevalues, but extending that frequency band to lower values ispractically not feasible.

At the output, a 4-bit programmablecapacitor bankprovides filtering.A pullup resistoris added to obtaingood linearity.

Biasing is done with a3-bit programmable on-chip voltage reference.

OUT

IN

VCB

and

gap

Simulated values for NF and IIP3 are 2.4 dB and −10dBm, at a maximum gain of 22 dB, with a powerconsumption of 12 mW.

Gain switching is achieved with a bypass cascodetransistor that diverts a part of the signal current tothe power supply for lower gain without influencingthe input matching

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RF

LO

B:1 1:B

RF+ RF-LO-

LO+ LO+

io+ io-

Wideband Down-Conversion Mixer

• The Gilbert cell is is used for wideband operation up to 6 GHz.

An NMOS input pair is used as a transconductance, driving RF signalcurrent into the core switch transistors that form the Gilbert cell

The folded switchingPMOSs can reduceflicker noise. Theextra foldingtransistors willcontribute a certainamount of thermalnoise, causing theoverall receiver’s NFto deteriorate.

The noise contributions in a switching mixer are not easy tounderstand or analyze but can generally be kept within limitsby using large LO signals and reduced dc current through theswitching transistors.

The switchable gain is achieved bydigitally programmable current gain B.

The input must be designedcarefully, as it will determineboth the noise and thelinearity performance of themixer.

A considerable biasingcurrent of 5 mA.

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Signal Selection and Dynamic Range

• How to handle: Signal Selection and Dynamic Range

• Signal Selection

Capturing a slice of bandwidth while rejecting adjacent frequencies, whichsometimes contain signals of higher power than the signal of interest.

• Dynamic Range

Defined by the max. and min. signal levels that the receiver can processwithout distortion that would degrade the SNR to an unacceptable level.

• Heterodyne Receivers

1. Convert an RF signal into an IF. The IF is passed through high-Q BPFs toremove undesired signals such as the image and interfering signals.

2. This IF approach works well for systems with defined channel bandwidths.But large banks of fixed filters would be required to cover the broad range ofpossible channels in SDR applications.This is not a practical solution.

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Variable-bandwidth Problem

• One possible solution: switched capacitor circuits.

The bandwidth is programmable by varying the capacitor ratio and the clockfrequency of the switched capacitor circuit.

• Another way to handle varying channel bandwidths is withthe use of direct conversion receivers (DCRs).

� Since no image frequency is produced, thus RF preselect filters can beeliminated.

� The removal of adjacent channel energy no longer requires high-Q BPFs butcan be accomplished with LPFs, which are much easier to integrate. This is agreat advantage because it is possible to integrate LPFs with programmablegain and bandwidth in today’s technology.

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Problems of DCRs

• Challenges: DC offsets and 1/f noise

• DC offsets are caused primarily by mismatch and/or by LOsignal coupling into the mixer RF port.

The undesired effect is saturation of the following dc-coupled gain stages.

• 1/f noise is the dominant source of noise in MOS transistors atfrequencies below100 kHz.

� In most CMOS processes, PMOS devices have between 2 and 5 times less 1/fnoise than do NMOS devices. Where this is true, PMOS devices should be usedin parts of the circuit where reducing 1/f noise is critical.

� Noise has a cumulative effect in a gain lineup, so the gain in the first stageshould be as large as possible.

� Chopper stabilization can greatly reduce the effects of any remaining 1/f noise.

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Transmitter Building Blocks

• The pre-power amplifier (PPA) is the final block in the SDRtransmit path.

The effective output power is not as high, and for manyapplications the average swing is much lower than the peak.Furthermore, a non-negligible voltage drop across the seriesresistance of the inductor sets the dc output voltage belowthe power supply.

INN

INP

OUT

The PPA includes extensiveprogrammability of gain settings.

The output stage is an inductively loaded CSamplifier with programmable bias current foroptimal linearity vs. power trade-off.

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DCFB

IN

OUT

Ref

Programmable Gain PPA

• The pre-power amplifier provides gain programmability.

The core of the amplifier is a CSstage with a PMOS resistive load.

3 additional PMOS transistors are placed in parallel with themain load to control the gain. Their gates can be connected toVdd , to turn them off and increase the gain, or to ground, toput them in the linear region and decrease the gain.

Changing the resistive load has animpact on the dc voltage and so, on itslinearity.

A dc level feedback circuit(DCFB) controls gate bias (forenhance linearity).

The total bias currentthrough the amplifiercan be controlled tooptimize the powerconsumption for thelinearity required.

The performance of thiscircuit varies widely, ofcourse, over carrierfrequency, required outputpower, bias and gainsettings, and so on.Simulation results indicatea total gain range of 50 dBand typical IM3 distortionlevels of −35 dB at 0-dBmoutput power.

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Direct Conversion Transmitters (I)

• BB digital process provides complex samples of the encodingintended, including encryption, pulse shaping, andlinearization preconditioning. These discrete samples arelowpass-filtered and amplified by the post-baseband (PBB)amplifier.

AmpLPF

AmpLPF

RFPowerAmp

DifferentialQuadrature

LO

I

Q

Bas

e B

and

dig

ital p

roce

ssin

g&

Dig

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o-A

nalo

g co

nver

ter

0 °

90 °

Frequency

dBm

leve

l OccupiedSignal

Bandwith

Far OutNoise Level

dBc/Hz

Carrier frequency

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Direct Conversion Transmitters (II)

• The benefit of this direct-launch Cartesian encoded carrier ismulti-mode compatibility with baseband frequency bandwidthand mask determination.

This enables additional digital processing technology to be applied through theentire transmitter, such as feedforward or predistortion linearization.

• Any form of amplitude, angle, frequency, or any combinationof modulation formats and bandwidths with no exceptions,including complex non-continuous multi-channel signals.

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Direct Conversion Transmitters (III)

• Three main issues:

� Sideband noise level

� Local oscillator feedthrough

� Self-generated interference (LO pulling causes remodulation of the carrier)

• Sideband noise level

In practice, it is introduced when circuit noise is increased in level by thebroadband gain in the transmitter system.

• Local oscillator feedthrough

The transmission gate ring switching mixer can improve carrier feedthrough. Withcareful design of the mixer, carrier feedthrough of better than −50 dBc is attainable.

• LO pulling

Shielding, grounding, .etc.

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Far-out Sideband Noise Contribution

• Far-out sideband noise canbecome an interference toreceivers close to thetransmission signal. (ForGSM, a BPF is often used toreduce the far-out noisesignificantly outside TX bands.)

• The lack of broadbandtunable RF bandpass filtersresults in far-out noise overa very wide range offrequencies and the TXmayoffend a receiver in closeproximity.

AmpLPF

AmpLPF

PADifferentialQuadrature

LO

I

Q

Bas

e B

and

dig

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roce

ssin

g&

Dig

ital-t

o-A

nal

og

co

nve

rter

0 °

90 °Gain = G

Low Pass FilterReduces Far Our noise contribution

Noise Figure contributes toInput Referenced Added Noise

Transmission GateSwitching Mixers

InputReferenced

Added Noise

CarrierVCO

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LO Frequency Pulling

• DCTs have an output frequency equal to that of the oscillatorsignal source frequency.

The interference causes perturbations in the VCO that are not corrected by the PLLcontrol loop. This undesirable remodulation of the VCO signal frequency willdegrade the quality of the signal transmitted.

AmpLPF

AmpLPF

PADifferentialQuadrature

LO

I

QBas

e B

and

dig

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g&

Dig

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o-A

nalo

g co

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ter

0 °

90 °

Transmitter radiated signal coupled input VCO signal source

AntennaNetwork

DC supply and groundconducted into VCOsignal source network

Electromagnetic Shielding

CarrierVCO

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Remodulationfrom Pulling

• Overcome Remodulation:

� Shielding

� Grounding

� Decreased VCO sensitivity to electromagnetic signals and the use ofsubharmonic, higher harmonic, or translated reference signal frequency.

� Lowering the inductive and capacitive coupling coefficient of the VCO resonantnetwork, use of differential VCO, and integrated implementation with a lowercoupling area profile.

• These are combined with multiple layers of isolation shieldingbetween the antenna and the VCOto form a remodulationrejection system.

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Required Isolation in ICs

Substrate

modulator PA

LO

ωLO

paraC

PLLsynthesizer

Isolation > 90~110 dB

PA < 30 dBm (1W)LO < 10 dBm

In the experiment, LO phase noise degrades when the injection power is as low as −80 dBm.

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Substrate Coupling and Isolation

• Dominant

� Diffusion capacitive coupling

� Impact ionization

� Inductive coupling (power grid fluctuations)

• Less significant

� Gate-induced drain leakage (GIDL)

� Photon-induced reverse current

� Diode junction leakage current

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Diffusion Capacitive Coupling

•

� SPICE models such elements with "CJ0" and "CJSW" (source/drain-to-substrate capacitance).

• Metal-to-metal capacitors:

Largest parasitic capacitance to the substrate, hence if these devices are used forimplementing large on-chip capacitors, they can act as significant substrate noise-injectors.

• As technology feature size reduces,higher doping concentration leadsto higher depletion capacitance andhence more coupling effects.

( )1 2

1 22jsb

q N NC

V N N

εψ

= + +

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Impact Ionization

• Reduced transistor feature sizes increase the electric field in thechannel and therefore impact ionization currents are becoming moresignificant compared to other injection mechanisms.

• In saturation, impact ionization takes place with a high electric fieldin the depleted region. For a p-type substrate, the generated holesare swept to the substrate generating an effective drain-to-substratecurrent. Recent experimental evidence suggests that hot-electroninduced substrate currents are the dominant cause of substrate noisein NMOSFETs up to at least one hundred megahertz.

• Shorter device channel lengths in advanced technologies are likelyto increase the impact ionization currents due to increasedchannelfields and smaller oxide thickness and drain junction depth.

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Inductive Coupling - Power Grid Fluctuations

• Due to parasitic effects (mainly bond wire inductance), powersupply lines become very noisy because of currents drawn bythe switching digital circuits. These currents induce largevoltage glitches when they switch (Ldi/dt noise) at substrateand well contacts. In addition, the power grid noise can be alsocapacitively coupled through metal-to-substrate parasiticcapacitance.

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Isolation in Silicon Substrate (Baseline)

• The baseline:D = 120 µmIsolation ~ 29 dB

� As the frequency increases, theisolation is getting more and moreworse.

� In the following slides, some isolationtechniques were applied to compare theisolations.

Baseline Isolation

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Isolation Techniques – P+ Guard Ring

• P+ ring: D = 120 µmW = 3 µm / d = 10 µmIsolation ~ 65 dB

• Guard rings sink the noise currents

P+ ring w/ low ohmic contact Isolation w/ P+-ring

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Isolation Techniques – N+ Guard Ring

• N+ ring:D = 120 µmW = 3 µm / d = 10 µmIsolation ~ 65 dB

• Good isolation at lowfrequencies due to the high capacitiveimpedance of the p-n junction between n-ring and p-sub.

Isolation w/ N+-ring

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Isolation Techniques – Deep N-Well Ring

• DNW ring:D = 120 µmW = 3 µm / d = 10 µmIsolation ~ 90 – 70 dB

• Good isolation at lowfreq.

• The DNW isolationdegrades with increasingthe frequency slower thanthe n-well due to thatDNW is lightly dopedthan the regular n+ well.(p-n junction capacitance with the psubstrate is smaller than that of then+ well)

Isolation w/ DNW N+-ring

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Isolation Techniques – Deep Trench

• Deep trench:D = 120 µmW = 3 µm / d = 10 µmIsolation ~ 65 dB

• A deep trench is a trench in thesilicon substrate approximately10 µm deep that is filled withoxide.

• The oxide in the deep trenchacts as a high impedanceinsulator that forces thesubstrate noise current to divedeep in the substrate.

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Isolation v.s. Distance D – Baseline

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Isolation v.s. D – P+ Guard Ring

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Isolation v.s. Ring Enclosure d

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Isolation v.s. Ring Enclosure Width w

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Adopt the Offset PLL

• Using an offset PLL eliminates remodulation by increasing thePLL loop bandwidth to include the signal modulation radiated.

0 °90 °

Quad Generator

ReferencePhaseFrequencyDetector

Narrow bandLPF Synthesizer

control

Offset VCO

Ft or 2*Ft

F = Ft or 2*Ft +/− Ft

Fr

VeryWideband

LPF

CarrierVCO

control

Ft or 2*Ft

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Using DDS

• An alternative direct-launch architecture that does not have aVCO operating at the transmitter output frequency uses directdigital signal synthesis (DDS).

• Briefly stated, a high-frequency clocking signal is used togenerate an output signal source without the use of anoscillator operating at the output frequency. DDS can be an all-band signal source for SDR, assuming that it is technicallypractical for up to 6-GHz operation with acceptable powerdissipation.

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Cartesian and Polar Implementation

• The advantage of polar processing is direct phase encoding onthe signal source and the potential of an efficiencyenhancement implementation in the amplitude modulation.

� First, the bandwidth associated with the magnitude and phase signals are multiplesof the Cartesian equivalent signals.

� Second, the magnitude and phase terms are very different, which causes divergentsignal processing effects, including potential time alignment imperfections.Thesebandwidth and time alignment effects require additional complexity andconsideration for each mode of operation.

Quadphase - Q

QMagitude =

( ) ( ) 1/22 2= I t +Q t

InPhase - I

Cartesian

( )A t

( ) ( ) ( )Phase = t = arctan Q t /I tθ ( )v t

Q

90°

-90°

180°

( )A t( )tθ

( )v t

I0°

Polar

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RF Power Amplification (I)

• Power Sources:

� Portable equipment: < 4 W of peak power (3.6- to 9-V dc battery)

� Fixed-base station equipment: < 100 W of peak power (110-V ac source)

� Vehicle mobile class: 10-W or higher (12-V dc battery)

• As the number of modes and bands increases, the size, cost,and performance of a bank of single solutions will becomeprohibitive without technology advancement in the passivefrequency-determining elements needed for matching at eachport of the power transfer gain stages within the transmittersystem.

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RF Power Amplification (II)

• Combining bands to expand the RF PAacross applicationswith an acceptable compromise of optimized single-solutionperformance.

• Multi-mode operation would be divided between TDDor FDD.

• The TDDswitches between transmitter and receiver operationmodes, where only one is active at any given time.

AmpLPF

AmpLPF

DifferentialQuadrature

Bas

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and

digi

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Dig

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I

Q

0º

90º

PA

Tx RxF = F

Mulitiplexer

AmpLPF

RxFDifferentialQuadrature

LNA

0º

90º

AmpLPF

I

Q Bas

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and

dig

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ssin

g&

Ana

log-

to-

Dig

ital c

onv

erte

r

TxF

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RF Power Amplification (III)

• The FDDhas simultaneous transmitter and receiver operationat different frequencies. Another incremental step would useprogrammable frequency-matching or device- operatingconditions to expand the band or mode of applicationassociated with a common RF power amplifier implementation.

AmpLPF

AmpLPF

DifferentialQuadrature

Bas

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and

digi

tal p

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ssin

g&

Dig

ital-t

o-A

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nver

ter

I

Q

0º

90º

PA

Tx RxF = F

Duplexer

AmpLPF

RxFDifferentialQuadrature

LNA

0º

90º

AmpLPF

I

Q Bas

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and

digi

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ssin

g&

Ana

log-

to-

Dig

ital c

onve

rter

TxF

Department of Electronic Engineering, NTUT51/82

RF Power Amplification (IV)

• The RF signal levels associated with power amplifiers canreach levels greater than the dc voltage source used forprogramcontrol.

• If the RF signal voltage is greater than that of the dc controlrange of the varactor, its capacitance is a function of the RFsignal in combination with the dc control voltage.

The capacitance is a function of the RF signal and is no longer constant at all times.This can result in modulation of the programmable element’s impedance value anddistortion in the RF signals.

Department of Electronic Engineering, NTUT52/82

RF Power Amplification (V)

• A single RF PAwould be an RF PAwith continuous operationacross all frequencies and optimized power dissipation atevery multi-mode format across all operating power levels. Inaddition, it would provide acceptable linear performance andharmonic content for any mode or band of operation.

• As the bandwidth approaches from100 MHz to 6 GHz, acascade of wide bandwidth-matching network gain stagesbecomes a complex design. Programmable elements couldbecome part of the solution to provide a design that wouldcompete successfully with a band of multi-mode or band-optimized performance solutions.

• However, a vector combined distributed implementation(distributed amplifier) can provide wideband performance witha modest increase in complexity.

Department of Electronic Engineering, NTUT53/82

Transmitter Efficiency

• A linear class B amplifier peak power occurs ideally when thepeak output voltage is equal to that of the dc supply voltage:

• A modulation format with a peak-to-average ratio of 3 dBwould have an efficiency reduction by a factor of 2.

• Supply Modulation:

Modulating the dc supply voltage as a function of the encoded output powermagnitude to provide peak operating efficiency at all power levels. There are anumber of supply modulation variations, such as envelope following, envelopeelimination and restoration, and polar modulation format.

0.7854 4

out out

in dc

P V

P V

π πη = = = =

Department of Electronic Engineering, NTUT54/82

Transmitter Linearization

• Error vector magnitude (EVM) and adjacent channel powerratio (ACPR) are used to measure the performance.

• ACPR:

A measure of the unintended generation of electromagnetic energy from thetransmitter network into frequency bands adjacent to the intended operatingfrequency band.

• A transmitter frequency-domain mask is defined by standardsto limit transmitter adjacent channel interference signal level.The source of the unwanted adjacent channel energy isnonlinear distortion products within the RF power amplifier.

Department of Electronic Engineering, NTUT55/82

Reducing Nonlinear Distortion

• Reducing the output power reduces the ACPR at the expenseof RF power amplifier efficiency, which is not always apractical solution.

• Linearization adds complexity to the amplifier to compensateor cancel generation of the distortion components.

Linearization Tech. Advantages Disadvantages

Feedback35 dB improvementNot depend on carrier frequency

Narrowband, < 1 MHzInstability potentialDependent on output termination

Feed-forward35 dB improvementNot depend on carrier frequencyWideband < 100 MHz

Efficiency performanceComplexity

PredistortionReduced hardwareTable-basedWideband < 10 MHz

Limited improvement (~15 dB)

Department of Electronic Engineering, NTUT56/82

Transmitter Stability

• When a circuit becomes unstable, it produces output signals atfrequencies not represented within the input signal.

The most common form of unstable behavior in a circuit is positive feedback, wherethe output is coupled to the input at a level higher than the original input signal. Asthe bandwidth of operation increases, these unintended feedback paths becomemore difficult to avoid.

• When an RF power amplifier is working, the operatingcondition is changing as a function of the input signal level.Therefore, the s-parameters are time-varying functions,making complete stability analysis more difficult to define.

Department of Electronic Engineering, NTUT57/82

Broadband LO Generation

• Perhaps the greatest challenge for SDRs is the generation ofLO oscillator signals over broad, continuous frequency ranges.

• Given that direct conversion is the receiver topology of choice,quadrature generation must be part of the solution. This is anarea that requires innovation to achieve SDR performance.

Department of Electronic Engineering, NTUT58/82

Phase-Locked Loops (I)

• PLLs are the most common implementations of frequencysynthesizers. A VCOis phase-locked to a stable referencefrequency (typically, a xtal) through a feedback path.

VCO architectures: Ring oscillators have found applications in some wirelessLAN standards and broadband TV tuner applications. However, the LC-tunedoscillator is most commonly used in applications with stringent phase noiserequirements.

• In addition to the benefits of the PLL integration into CMOSprocesses, they typically exhibit good performance in terms ofphase noise, current drain, area, and spurious performance.

The greatest challenge for PLLs is making them tunable over a broadfrequency range. Tuning ranges of 20% are typical but can be as high as 30%.Recent results suggest that ranges approaching 50% are possible.

Department of Electronic Engineering, NTUT59/82

Phase-Locked Loops (II)

• Even a 50% tuning range is not enough, there are a multitudeof approaches for extending the frequency range of PLLsystems. One approach is to have multiple VCOs, which inmost cases can share the same PLL circuitry.

• The size of inductors places a practical limit on the number ofVCOs and therefore the achievable frequency range. Whetheron- or off-chip, cost and manufacturing issues prevent the useof huge banks of VCOs to cover wide tuning ranges.

• Another way to get a broad tuning range is to have one ormore VCOs and multipliers, dividers, or mixers combinations.

Inevitably, the addition of these circuits comes at the cost of higher powerconsumption for equivalent noise and spurious performance.

Department of Electronic Engineering, NTUT60/82

Direct Digital Synthesizer (DDS)

• DDSs have a broad frequency tuning range, fine frequencyresolution, and very fast switching

• ROM consumes a large amount of power.

• Synthesized frequency is lower than the reference frequency.

Acc. Output

ACC

n mLPF outF

refF

DACROMROM Output DAC Output

Acc

. O

utp

ut

RO

M O

utp

ut

DA

C O

utp

ut

outF

Department of Electronic Engineering, NTUT61/82

Digital-to-Time Converter

• Use of a digital-to-time converter (DTC) to construct an outputfrequency fromthe phase information in the accumulator tolower power consumption.

Acc. Output

ACC

Kn m

Tap0TapX

PhaseDetector

ChargePump

LPF

vtuneTapped Delay Line

Tap0 Tap1Tap2…… TapX-1 TapX

Tap Selection Logic

……

Digital-to-Time Converter

outF

refF

Department of Electronic Engineering, NTUT62/82

Example (I)

• A highly reconfigurable low-power transceiver implemented ina 90-nmCMOS process.

• The RFIC processes signals of multiple wireless protocolsfrom 100 MHz to 2.5 GHz with −6 dBmand a voltage gain of48 dB.

• The transmitter has better than 40 dB of carrier suppression,35 dB of sideband suppression, and an EVMof 1% at 800MHz.

• The frequency synthesizer uses direct digital synthesis toachieve instantaneous frequency switching and a phase noiseof −115 dBc/Hz at 25 kHz offset for a 500-MHz carrierfrequency.

Department of Electronic Engineering, NTUT63/82

Rx DDS ReferencePLL/VCO

PMA VGA BQ

DCOC MUX

Tx TEST POINTS

SPI

ButterworthForward pole 2 Input BufferTx Forward DDS

ReferenceTx Reverse DDSCartesian Rev Amp/Mixer

Cartesian BB Forward

Reference

Example (II)

Programmable LPF, 4 kHz ~ 10 MHzDDS

DDS

DDS

PLL

DCR

DCT

Cal.

Cal.

external connection toan ADC and digitalprocessing.

external connection toan DAC and digitalprocessing.

Programmable LPF, 4 kHz ~ 10 MHz, 10% BW

step

90 dB VGA powercontrol

For external

LNA

Chopping

Department of Electronic Engineering, NTUT64/82

QuadradtureMixers

• The quadrature mixers on RXinputs 1, 3, and 5 arenonchoppedpassivemixers built with a quad ring of CMOStransmission gates.

• The quadrature mixers on RXinputs 2 and 4 use dynamicmatching to improve IP2, flicker noise, and dc offset. Thechopping mixers are built with three mixers in, where eachmixer is built with a quad ring of CMOS transmission gates.

RFp

LOm OUTp LOp OUTm

RFm

RFp

RFm

CH

OP

p

CH

OP

m

LO

p

LOm

OUTm

OUTp

Department of Electronic Engineering, NTUT65/82

Baseband Filters (I)

• Baseband filters that support multiple bandwidths areimplemented along with gain control and dc offset correction.

• A filter bandwidth is programmable from4 kHz to 10 MHz in6.25% steps or less. Sufficient margin is built into the designto allow for a 20% change in RC tolerance and still maintainthe bandwidth range of 4 kHz to 10 MHz.

• Bandwidth selection is implemented by adjusting the resistorand capacitor values in the filter design.

Department of Electronic Engineering, NTUT66/82

Baseband Filters (II)

• Baseband filter gain control is accomplished at three points:

� A programmable resistor divider at the input of the PMA allows attenuation infour 6-dB steps.

� The PMA has a maximum gain of 32 dB and a minimum gain of −10 dB.

� The VGA has a gain range of 8 dB, and the output buffer has a programmablegain control of 0 to 18 dB in 6-dB steps.

� The entire baseband filter lineup has a maximum gain of 64 dB and a minimumgain of −4 dB.

• A notable feature of the baseband filter is the use of chopperstabilization to mitigate the undesirable effects that occur indirect-conversion receivers when designed in a CMOS process.

Department of Electronic Engineering, NTUT67/82

Chopper Stabilization

• Chopper stabilization isimplemented around the first-stageamplifier of each two-stage op-amp(OPA’s input-referred voltage offsetand flicker noise performance areheavily dependent on the first stageof the OPA).

• The chopper frequency is derivedfrom the crystal input and can beselected as a divide by 1, 2, 4, or 8of the crystal frequency. Flickernoise is essentially eliminated whenchopping is enabled, and thusnarrowband protocols will seeimprovement in receiver sensitivity.

Department of Electronic Engineering, NTUT68/82

DC Offset Correction

• Dc offset correction circuitry (DCOC) is implemented as acomplete control loop that corrects dc offsets automatically atthe output of the baseband filter.

• DCOC consists of a 1-bit ADC (comparator), control logic,and a 5-bit current-mode DAC that injects current into thefeedback resistors of the VGAto adjust the offset voltage.

• The control logic implements a successive approximationalgorithm that converges on the correct 5-bit word thatcompensates for the filter’s dc offset.

Department of Electronic Engineering, NTUT69/82

Performance of This Transceiver

Summary of the Transceiver Performance

Frequency range 100 MHz ~ 2.5 GHz

RX NF 7 dB

RX gain 48 dB

RX IIP2 +60 dBm

RX IIP3 -6 dBm

RX current 40 mA

TX output power +6 dBm

TX sideband suppression 35 dBc

TX current 40~90 mA

EVM (pi/4 DQPSK 3.5 Msps) 1%@800 MHz

LO phase noise −115 dBc/Hz@25 kHz

LO frequency resolution 15 Hz

LO current per DDS 80 mA

Department of Electronic Engineering, NTUT70/82

Adaptive Multi-mode RF Front-ends

• To provide various services fromdifferent wirelesscommunication standards with high capacities and high datarates, integrated multi-functional wireless devices are required.

• In the current multi-standard scenario, transceivers are mostlyimplemented by replicating the radio-frequency (RF) front endfor each operating standard and by sharing partially the analogbaseband circuitry, but with a number of additional switches.

Although this approach allows for an optimal performance optimization across thebands, the increase in hardware required to implement such a multifunctionalwireless device increases the total silicon area and cost and may reduce the usetime compared to single-standard implementations.

Department of Electronic Engineering, NTUT71/82

Sharing Building Blocks

• By sharing building blocks between different applications andstandards, portable wireless devices potentially gain advantageover existing devices:

� They use a smaller chip area and have a potential for lower overall cost. Thisrequires the development of adaptive circuits and systems that are able to trade offpower consumption for performance on the fly.

� Realization of adaptivity functions requires scaling of current consumption to thedemands of the signal processing task.

Department of Electronic Engineering, NTUT72/82

Adaptive Multi-mode Low-power Design

• Mobile wireless equipment today is shaped by user andapplication demands and RF microelectronics.

• Main drivers for mobile wireless devices are related to cost,which depends on volume of production, size of mobile units,engineering bill of materials, power consumption, andperformance.

� Power consumption depends on available frequency spectrum, functionality, andperformance.

� Performance depends on applications, standards, and protocols. Wireless systemsfor new applications require an extension of the capabilities for the RF devices oftoday, creating an opportunity for low-power adaptive and multifunctional RF ICs.

Department of Electronic Engineering, NTUT73/82

Low-power and Adaptive RF Circuit

• A variety of applications:

Supporting the transfer of text, audio, graphic, and video data, maintainingconnection with many other devices, position aware, and perhaps wearable.

• A combination of multiple functional requirements and limitedenergy supply froma battery is an argument for the design ofboth adaptive low-power hardware and software.

• An adaptive design approach poses unique challenges:

From hardware design to application software, and ultimately throughout all layersof the underlying communication protocol.

Department of Electronic Engineering, NTUT74/82

Adaptive Topology

adaptive analogRF front-end

LNA

I-mixer

Q-mixer

Quadraturegeneration

VCO

VGA

VGA

A/D

A/D

DSP

Memory

CPU

adaptive analog base-band adaptive digitalback-end

( )x t

( )y t

( )I t

( )Q t

Setting the performance parameters of an RF front end by means of adaptive circuitry is a wayto manage power consumption in the RF path of a receiver. An adaptive LNA, an adaptivemixer, and an adaptive VCO allow more efficient use of scarcebattery resources.

Furthermore, adaptive analog baseband and digital back-endcircuits enable complete hardware adaptivity (monitoringandadjust TRX parameters).

Department of Electronic Engineering, NTUT75/82

Multi-mode and Adaptive RF Circuit (I)

• GSM, UMTS, Bluetooth, 802.11a/b/g, GPS, and DVB-H aresome of the standards likely to be present in the multi-standard,multi-mode, multi-band mobile terminals of the future.

• The design of multi-functional wireless devices isaccompanied by various challenges:

� System challenges:

Single high-performance, low-power terminal that is cheaper than acompound of separate single-mode terminals.

� Circuit design challenges:

For full integration of multifunctional devices include on-chip imagerejection and provision of wide bandwidth and dynamic range.Technological challenges include the integration of low-cost and high-performance-scaled silicon devices on a chip.

Department of Electronic Engineering, NTUT76/82

Multi-mode and Adaptive RF Circuit (II)

• Multi-standard modules can be implemented in various ways:

� As stand-alone circuits that are designed for the worst-case condition of themost demanding standard.

� As multiple circuits (i.e., one per standard).Even though simpler to implement,this approach requires more silicon area. Moreover, when multiple standardsoperate simultaneously, power consumption increases.

As stand-alone adaptive circuits. When different standards do not operatesimultaneously, circuit blocks of a multi-mode handset can be beneficiary shared,offering power and area savings compared to other multi-standard receiverimplementations, such as multi-standard receivers implemented using circuitsdesigned for the worst-case condition and multi-standard receivers implementedwith one receiver circuit per standard.

Department of Electronic Engineering, NTUT77/82

Multi-mode and Adaptive RF Circuit (III)

• For adaptive LNAs and mixers, power consumption is tradedoff for dynamic range, whereas adaptive oscillators trade offpower consumption for phase noise and oscillation frequency.

• After a signal is down-converted to the baseband, it is filtered,amplified, and digitized.

� To accommodate multiple radio standards with different bandwidths andmodulation schemes, multi-mode LPFs need to compromise bandwidth, centerfrequency, selectivity, and group delay for optimal dynamic range and powerconsumption.

� Multi-mode ADCs have to sample signals belonging to different standards,tailoring different sample rates, dynamic range, and linearity requirements.

Department of Electronic Engineering, NTUT78/82

Multi-mode Receiver Concept (I)

• Wireless devices may use a common receiver if the protocolsof the radio standards support intersystemoperability.

DCS1800

LNA

LNA

LNA

WCDMA

WLAN,DECTBluetooth

MMA-QD IC

I-IF

Q-IF

VCO

buffer

buffer

2-stagePolyphase

filter

Differentialamplifier

Differentialamplifier

I-mixer

Q-mixer

Impedance matching, packaging,and prefiltering requirementsare relaxed and simplified byusing multiple LNAs.

An RF switch selectsthe mode of interest.

If the VCO and mixerperformance is adequateto cover the range ofsignals anticipated foreach application, thequadrature downconverterenables a multi-standardreceiver realization with asingle circuit block(multi-mode adaptivequadrature down-converter, MMA-QD IC).

Department of Electronic Engineering, NTUT79/82

Multi-mode Receiver Concept (II)

• Adaptivity toRequirements for Various Standards Referred to LNA Input

DCS1800 WCDMA WLAN Bluetooth DECT

f0 (GHz) 1.8 2.1 2.4 2.4 2.4

NF (dB) 9 6 10 23 18

IIP3 (dBm) −9 −9 −12 −16 −20

PN@1MHz (dBc/Hz) −123 −110 −110 −110 −100

Receiver Noise and Linearity Specification per Mode of Operation

Specification Demanding modeDCS1800/WCDMA

Moderate modeWLAN 802.11b

Relaxed modeBT/DECT

NF (dB) 6 10 18

IIP3 (dBm) −9 −19 −16

PN@1MHz (dBc/Hz) −123 −110 −100

Department of Electronic Engineering, NTUT80/82

Multi-mode Receiver Concept (III)

• Referring to the channel spacing of the standards considered,zero-IF and low-IF receiver configurations may be supported.For example, the multi-mode adaptive quadrature down-converter may operate in zero-IF mode for all standardsconsidered except the 200-kHz narrowband GSM(DCS1800band) standard where low-IF operation would be favored (dueto flicker nosie).

• The standards considered are chosen to illustrate the feasibilityof the adaptivity design concept for multi-mode receivers.

Department of Electronic Engineering, NTUT81/82

Multi-mode Receiver Concept (IV)

• Given the multi-mode receiver requirements and thespecifications for the LNAand baseband circuitry, the noisefigure and linearity performance of the quadrature down-converter can be determined for each mode of operation usingthe cascaded NF and IIP3 formulas.

The demanding-mode performance is met with a 0-dB gain of the down-converterwith an accompanying NF tuning range (NFTR) of 14.6 dB and an IIP3 tuningrange (IIP3TR) of 7.6 dB. Note that by trading off the performance of the LNA andbaseband circuitry, a different (more relaxed or more demanding) set of down-converter requirements results.

Required Performance for the Multi-mode Quad-Downconverter for 0 dB of Gain

Specification Demanding modeDCS1800/WCDMA

Moderate modeWLAN 802.11b

Relaxed modeBT/DECT

NFqd (dB) 12.7 19.75 28.8

IIP3qd (dBm) 6.74 3.35 −0.87

Department of Electronic Engineering, NTUT82/82