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IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007 1749
Dual-Frequency Microwave Moisture Sensor Basedon Circular Microstrip Antenna
Mohamed Mustafa Ghretli, Kaida Khalid, Ionel Valeriu Grozescu, Mohd. Hamami Sahri, and Zulkifly Abbas
AbstractA dual-frequency sensor was developed to measure
moisture content of lossy liquids. The experiment is based on mea-
surements of far-field reflection at two frequencies in the X-band,8.48 and 10.69 GHz. The replacement of the conventional open
horn antenna with microstrip radiating patches makes the sensor
more versatile and compact. The sensor is interfaced to a note-
book computer using a multipurpose data acquisition board and
few custom made circuits. Data collection, errors correction and
calibration modules were written using LabVIEW programming
language. Three calibration methods were used to obtain an em-
pirical equation, which is applied to obtain the moisture content
of the rubber latex samples. Moisture content can be determined
with a standard error 0.49%.
Index TermsDual microwave frequency sensor, microstrip an-tenna, moisture content, rubber latex.
I. INTRODUCTION
IN THE LAST few decades, microwave dielectric-based sen-
sors were developed for moisture or water content determi-
nation in different materials [1]. The fact that these sensors rely
on measurement of dielectric properties, which are also depen-
dent on other physical properties of the material, i.e., bulk den-
sity, moisture content, temperature, etc., makes them potential
multiparameter sensors provided that appropriate correlationsare established between the measured dielectric properties and
the physical properties of interest [2].
Temperature variation causes errors in all indirect moisture
measurement methods. A common method to eliminate tem-
perature errors employed in industry involves measurement
the sample temperature and compensates for temperature
variations. To increase the accuracy of moisture content de-
termination and to reduce the influence of temperature, a
two-parameter measurement can be used. For example, in
many cases, using the magnitude ratio of reflected waves at two
microwave frequencies in X-band region (8.48 and 10.69 GHz)
Manuscript received January 22, 2007; revised August 16, 2007; acceptedSeptember 3, 2007. This work was supported in part by the Ministry of Sci-ence, Technology and Innovation, Malaysia, and in part by the University PutraMalaysia, Selangor, Malaysia. The associate editor coordinating the review ofthis paper and approving it for publication was Prof. Robert Trew.
M. M. Ghretli was with the Department of Physics, University PutraMalaysia, Serdang, Selangor, Malaysia. He is now with the High Institute ofComputer Technology, Tripoli, Libya (e-mail: [email protected]).
K. Khalid, I. V. Grozescu, and Z. Abbas are with the Department of Physics,University Putra Malaysia, Serdang, Selangor, Malaysia (e-mail: [email protected]; [email protected], [email protected]).
M. H. Sahri is with Faculty of Forestry, University Putra Malaysia, 43400UPM, Serdang, Selangor, Malaysia (e-mail:[email protected]).
Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/JSEN.2007.908920
Fig. 1. View of the instrument.
as a function of temperature, it is possible to obtain a tempera-ture independent behavior. This kind of setup does not involve
phase measurement and the cost will be drastically reduced.
The principle of the sensor is based on measuring reflection
of the mature fields in the far -field region from the sample
interface. This new computer controlled dual-frequency mois-
ture meter is developed for special applications, where the water
content of various lossy liquids and food products is an impor-
tant process parameter. A predetermined calibration equation
is used, avoiding the use of the expensive vector network ana-
lyzers. Three calibration schemes are described.
In this paper, we deviate from the open-ended coaxial line as
a common sensor for moisture measurements and we introducean alternative approach that uses a disk patch microstrip antenna
operating at two different frequencies in the X-Band at 8.48 and
10.69 GHz.
The advantages of the dual-frequency sensor in terms of mea-
surements accuracy are discussed as well. This sensor was tested
with Hevea rubber latex.
Although, the dual-frequency sensor was developed for tem-
perature independent measurements of the moisture content,
it can be operated in single-frequency mode when necessary.
Here, we focus in the design and the development of the dual-
frequency sensor only. An application of the sensor for temper-
ature independent measurement of the moisture content will be
presented somewhere else.
II. DEVELOPMENT OF MOISTURE METER
A general view of the dual-frequency microwave moisture
meter for the lossy liquids is shown in Fig. 1(a). In Fig. 1(b), the
circular microstrip antennas became visible. The test liquid con-
tainer is placed on the top of the microstrip antennas assembly.
The sensor system is interface to a notebook computer
through a data acquisition board (PCI NI-6024-E) having a
maximum sampling rate of 200 Ksamples/s and a 12 bit reso-
lution. This multipurpose board has 16 analog input channels
and 8 digital I/O lines. Six analog input channels are used
in a differential mode to measure the negative output signals
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1750 IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007
Fig. 2. Schematic diagram of microwave reflection system principle.
obtained from the two Schottky detectors and to monitor the
regulated supply voltage. The software interface was written
using LabVIEW programming language. The software incor-
porates beside the data acquisition module, an error correction
module and a system calibration module.
A. Theory
In this work, the microwave reflection system uses a bistatic
air reflection method to determine the water content in rubber
latex samples. The method employs measurements of mi-
crowave reflection magnitudes, from the sample interface, in
a region where the angular field distribution of the antenna is
essentially independent of the distance from the antenna. This
region is known as the far-fields or Fraunhofer region [3]. If the
source has a maximum overall dimension D and the wavelength
, the far-field region is commonly taken to exist at distances
greater than 2D from the source.
Schematic diagram of the microwave reflection system isshown in the Fig. 2. An incident microwave originating from
a transmitter situated the bottom of the sample experiences
in a series of reflections and transmissions from and through
the interfaces of a multilayer system. The reflection signal
arriving at the microwave receiver, situated in the same plane
with the transmitter, consists of a series of reflections which
occur at all system interfaces (air-perspex, perspex-sample, and
sample-air).
The four-layer system (air-perspex-sample medium-air) can
be visualized as a cascaded combination of two port networks.
The total reflectance of the system can be obtained from the flow
graph shown in Fig. 3 using Masons nontouching loops rule [4].
The ratio of the complex wave amplitude at the nodes (b) and(a) can be obtained as
(1)
where
(2)
, and represent the thickness, absorption coefficient, and
reflection coefficient of the medium, respectively. The sub-script letters and corresponds to air, perspex, and sample
Fig. 3. Signal flow graph for the four-layer system.
Fig. 4. (a) Schematic diagram for the microwave sensor with connectors.(b) Top view of the sensor.
medium, respectively. Perspex is a synthetic polymer com-monly called acrylic glass. This thermoplastic and transparent
plastic is sold by the tradenames Plexiglas, Perspex, Plazcryl,
Limacryl, Acrylex, Acrylite, Acrylplast, Polycast, and Lucite.
One must note that all the propagation constants, reflection
coefficients and impedances shown in (1) and (2) are complex
variables.
B. Sensor Design
Fig. 4(a) shows schematic diagram of the dual-frequency
sensor consisting of four circular microstrip patch antennas.
Fig. 4(b) displays the top view of the sensor. Two patches (T1
and T2) serving as transmitters are fed by two dielectric res-onant oscillators operating at the nominal frequencies of 8.48
and 10.69 GHz, respectively. The other two opposite microstrip
patches (R1 and R2) serving as the receivers, are connected to
Schottky wideband coaxial power detectors (MDS 1087-S).
The basic circular antenna geometry comprises a thin con-
ducting patch of radius a on a dielectric substrate with the di-
electric constant and the thickness , backed by a ground
plane. The conducting patch and the ground plane form the elec-
tric walls of a cavity. The resonator is excited by a line current
through a coaxial probe fed from the ground plane at the dis-
tance from the centre of the circular patch. Fig. 5 shows the
schematic design of a single circular microstrip antenna.
The four antenna elements are fabricated on a 1.59-mm-thicksubstrate of polytetrafluoroethylene RT Duroid, supported by a
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GHRETLI et al.: DUAL-FREQUENCY MICROWAVE MOISTURE SENSOR BASED ON CIRCULAR MICROSTRIP ANTENNA 1751
Fig. 5. Schematic design for a single circular microstrip antenna.
5-mm-thick aluminum disk to add strength to the assembly and
to hold the SMA coaxial connectors to each element.
Based on the cavity model [5], the resonant frequencies of
TM modes of a circular patch antenna are given by
(3)
where is the speed of light in free space, is the th root
of the first derivative of the -order Bessel function .
Due to the geometrical consideration, the dominant mode
for our antenna elements is . Consequently, the reso-
nant frequency can be obtained from (3) by introducing
. This resonant frequency does not take into ac-
count the effect of fringing along the edge of the resonator.
This effect makes the patch to look electrically larger. In 1995,
Kumprasert et al. [6] have introduced a correction replacing the
actual radius with an effective radius given by
(4)
Therefore, the dominant resonant frequency of the mode
should be modified, as shown in (5)
(5)
The actual radius of the circular patch antenna corre-
sponding to a desired resonant frequency, in our case 8.48 and
10.69 GHz, can be numerically obtain from (5). Table I shows
the properties and design parameters of circular microstrip
antennas.
When designing a number of antenna elements on a single di-
electric substrate mutual coupling between the elements has to
be taken in consideration. The detector connected to an antenna
element can receive radiation from a neighboring radiating el-
ement even in the absence of a reflecting sample. The mutual
coupling between the antenna elements is discussed in the nextsection.
TABLE I
MATERIALS PROPERTIES AND DESIGN PARAMETERS OF CIRCULAR
MICROSTRIP ANTENNA OPERATING AT 8.46 AND 10.69 GHZ
Fig. 6. Measured mutual coupling between two coaxial-fed circular microstripantennas, for E-plane coupling (dotted line) and H-plane coupling (continuousline).
C. Antennas Mutual Coupling
In general, mutual coupling is primarily attributed to the
space waves fields. It is dependent on the separation distances
between antenna elements. Another important coupling factor
is the surface waves. Surface waves exist and propagate within
the dielectric, and their excitation is a function of the thickness
of the substrate. For thin substrates, the contribution of the
surface wave to the mutual coupling can be neglected [3].Fig. 6 shows E- and H-plane coupling coefficients as a func-
tion of the distance between the radiating edges for a circular
patch of 4 mm radius, 157.5 m thickness, dielectric constant
of 2.23, and resonant frequency of 10.69 GHz. It can be ob-
served that for an edge separation satisfying the E-
and H-coupling coefficients have close values. is the reso-
nant wavelength of the antenna. In the range the
H-plane coupling configuration have a much lower coupling co-
efficient than the E-plane coupling configuration. Increasing the
distances between the radiating elements will further reduce the
coupling factor, but this increases the overall size of the sensor.
E-plane coupling configuration was chosen for the present de-
sign. The edge separation chosen was 0.89 that correspondsto a mutual coupling of dB. It satisfies one of our design
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1752 IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007
Fig. 7. (a) S parameter (return loss) for the circular microstrip antenna T2with a resonant frequency of 8.388 GHz. (b)
S
parameter (return loss) for
circular microstrip antenna R2 with resonant frequency of 8.510 GHz.
requirements that no more than 2% of the radiated power is al-
lowed to be coupled. The separation distance of each pair is
different since each pair operates at different frequency. The
distances from the edges of the circular patches to the edge of
the substrate is about , as a normal practice by many re-
searchers to reduce the side loops of the antennas radiation pat-
tern and increase the gain.
D. S-Parameters Measurements for 8.48 GHz and 10.69 GHz
Antenna Pairs
The two ports of the Virtual Network Analyzer (VNA) are
connected to 8.48 GHz antenna pair denoted by T2 and R2 in
Fig. 4(b). Port 1 is connected to antenna T2 and Port 2 is con-nected to antenna R2. Marker 2 in Fig. 7(a) shows a peak return
loss of 15.06 dB at 8.388 GHz. For the intended frequency of
8.48 GHz, the return loss is 10.29 dB.
Using antenna T2 as a transmitter fed by 8.48 GHz oscillator,
the radiated power will beless by 6% thanif it is driven by 8.388
GHz oscillator. Similar, if antenna T2 is used as a receiver, a 6%
reduction of power is detected at antennas output port.
To optimize the performance, it is a good idea to fabricate theantenna element and test it for its resonant frequency before or-
dering the oscillator. Dielectric resonant oscillator (DRO) has a
mechanical tuning to adjust its output frequency within a speci-
fied range. Unfortunately, 8.388 GHz is not within the mechan-
ical tuning range of the 8.48 GHz DRO used and about 6% of
radiated output power is lost.
The parameter for antenna R2 is shown in Fig. 7(b).
The spectrum has a peak at 8.48 GHz with the return loss of
dB. This means that about 96% of the input power is ra-
diated and/or lost by the antenna and only 4% reflected back to
the source, being an improvement over the performance of T2.
Similar S-parameters measurements have been done for an-
tennas T1 and R1. The return loss at 10.69 GHz for T1 and R1was 7.06 and 15.2 dB, respectively [see Fig. 8(a) and (b)].
E. Sample Level
Another parameter to be considered is the liquid (sample)
height, (Fig. 1). It is desired that the microwave reflection
signal amplitude to be independent of the sample height. Equa-
tion (1) can be used to predict the minimum value of the sample
height. Fig. 9 shows the normalized reflection power of the mi-
crowave signal as function of height for a sample of distillated
water and different thickness of the perspex substrate (1, 3, 4,
and 5 mm). The perspex constitutes the bottom of the samples
container.Distilled water has the attenuation constant of 139 m
and skin depth of 7.19 mm. It can be calculated that for dis-
tillated water sample having the height of or 28.76 mm, the
propagating waves are attenuated in a ratio of 99.965% before
reaching the sample-air interface.
To ensure a total attenuation of the microwave signal through
the liquid sample, we use a minimum 50 mm height for the
rubber latex samples.
F. Sample Holder Plate Thickness
The bottom of the sample holder was sought to be a material
with very low attenuation constant for the microwave region ofinterest. Due to the dielectric (skin depth of 8 m) and mechan-
ical properties (rigidity and scratch resistance), the perspex is an
ideal material for the present design.
The thickness of the perspex layer, (Fig. 1) is another sig-
nificant parameter of the design.
The considerations about the perspex layer thickness are not
based on the total attenuation of the microwave signal through
the sample as seen in the case of the sample height. They are
based on the microwave phase change as propagates though the
perspex medium. The superposition of the incident and reflected
depends on the phase relation between them. Fig. 10 shows the
dependence of the microwave normalized reflected power of
perspex layer thickness, corresponding to a water column heightof 50 mm at 10.69 GHz.
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GHRETLI et al.: DUAL-FREQUENCY MICROWAVE MOISTURE SENSOR BASED ON CIRCULAR MICROSTRIP ANTENNA 1753
Fig. 8. (a)S
parameter (return loss) for circular microstrip antenna T1 withresonant frequency of 10.64 GHz. (b)
S
parameter (return loss) for circularmicrostrip antenna R1 with resonant frequency of 10.82 GHz.
The lowest ratio of the reflected power (36.3%) occurs for
perspex plate thickness of 5.1 mm corresponding to a quarter
wavelength of a wave traveling in perspex medium at a fre-
quency of 10.69 GHz. The highest ratio of the reflected power
corresponds to a half wavelength of the perspex plate (10.2 mm).
Based on the availability of the materials at this time, the per-
spex plate thickness was chosen to be 3 mm, which corresponds
to the reflected power level of dB.
G. Distance Between Antenna and the Bottom of the Sample
Holder
The distance from the antennas plane and the bottom of thesample holder (perspex) must be optimized for the maximum
Fig. 9. Normalized power of the microwave reflected signal versus watercolumn height for various perspex thicknesses.
Fig. 10. Normalized reflected power amplitude(continuous line) andreflectioncoefficient phase (dotted line) versus perspex plate thickness.
reflected power and to satisfy the far-field condition. This dis-
tance is represented in Fig. 2 by thickness of the air layer .
Fig. 11 shows experimental reflection power as a function of air
layer thickness, corresponding to the perspex layer of 3 mm and
the semi-infinite water layer at 10.69 GHz.
The position of first maxima (16.5 mm) was chosen as the
optimum distance between the sample holder bottom and the
antennas plane. This distance satisfies as well the far-field con-
dition that requires a distance bigger than 8.72 mm for the patch
antenna of radius 6.21 mm (8.48 GHz) and bigger than 6.65 mm
for the patch antenna of radius 4.83 mm (10.69 GHz).
H. Oscillators Warm-Up
To ensure the reliability of the measurements, the oscillators
must be warmed up for at least 4 min before any measurements
are performed. Fig. 12 shows the voltage output of the receivingpatches, while the oscillator warmed up.
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1754 IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007
Fig. 11. Experimental reflection power versus distance between sample inter-face and antenna corresponding to 3 mm perspex layer thickness and semi-infi-
nite water layer at 10.69 GHz.
Fig. 12. Output voltage of the receiving patches as the oscillators warm-up.
III. MEASUREMENTS
The sensor was tested using samples of diluted concentrated
rubber latex. The samples were divided into two groups. The
first group, consisting of 12 samples with moisture content (MC)
ranging from 38.7% to 95.0% was used to calibrate the sensor
system. The second group, with the same number of samples,
but with random moisture content was used for the test.
A. Single-Frequency Calibration
After the oscillators have stabilized their output, the signals
corresponding to open air and a sample of distillated water are
recorded. The output of the receivers is sampled at a rate of 1000
sample/s. A total of 10 000 points are recorded and averaged for
each measurement. This attenuated substantially the white noise
present in the system.
The power of the reflected microwave signal from the inter-
face of each diluted latex solution sample (Pi) is normalized bythe value corresponding to the distillated water sample (Pw). In
Fig. 13. Calibration curves for diluted Hevea rubber latex at 8.48 and10.69 GHz.
TABLE II
POLYNOMIAL COEFFICIENTS OF THE CALIBRATION CURVES FOR THE
MOISTURE CONTENT SENSOR AT 8.48 GHZ AND 10.69 GHZ
the following discussion, for convenience, the normalized re-
flected microwave power (P/Pw) is simply denoted by .
It has been found that a third-order polynomial, as shown in
(6), can describe accurately the moisture content-reflected mi-crowave power characteristic of the sensors for both frequencies
(6)
Fig. 11 shows calibration graphs for distillated Havea rubber
latex at 8.48 and 10.69 GHz. The polynomial coefficients in-
volved in (6) are shown in Table II together with the goodness
of fit .
B. Dual-Frequency Calibration and Measurements
Validity of the calibration equations at 8.48 and 10.69 GHz
for the moisture content sensor was examined with a series of
12 samples with MC unknown (Fig. 13). In the beginning, themoisture content of each sample was determined for both fre-
quencies using (6) with the corresponding polynomial coeffi-
cients shown in Table II. One should note that the MC of the
samples could be determined using only any one of the two fre-
quencies. Ideally, the two values of the MC for a given sample
obtained at 8.48 and 10.69 GHz should be identical.
In the next sections, three methods used to improve the accu-
racy of moisture content determination are described.
1) Average Method Prediction: The predicted moisture con-
tent can be slightly improved using the average of the moisture
contend at 8.48 GHz and 10.69 GHz , as shown
in (7)
(7)
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GHRETLI et al.: DUAL-FREQUENCY MICROWAVE MOISTURE SENSOR BASED ON CIRCULAR MICROSTRIP ANTENNA 1755
2) Weighted Average Method Prediction: The accuracy
of MC determination could be further improved by using a
weighted average of and instead of normal average
(8)
where the weighting coefficients and satisfy the following
constrain:
(9)
The weight coefficients and can be calculated from the
standard deviation of the power reflection measurements at 8.48
GHz and 10.69 GHz , as shown below
(10)
with
(11)
is the -sample at first frequency (8.48 GHz), is the cor-
responding average value, and is total number of recorded
samples.
3) Multivariate Regression Calibration Method: Another
method used for moisture content prediction is based on a
multivariate model. One must recall that the system uses two
reflected signals and at two different operating frequen-
cies. The signals and represent two dependent variables
in this analysis. The independent variable is the MC of the
sample, obtained using standard oven technique.
The MC can be predicted using both reflection measurementsand through a third-order polynomial expression shown
in (12)
(12)
For the below discussion, it is useful to introduce the following
vectors:
(13)
(14)
(15)(16)
where is the polynomial coefficients vector, and
are the vectors corresponding to the observations at 8.48 and
10.69 GHz, respectively, is the standard moisture content
vector and is total number of observations. The reflected
power matrix, can be constructed as
(17)
TABLE III
POLYNOMIAL COEFFICIENTS FOR THE REGRESSION MODEL
Fig. 14. Residue comparison for the MC determination methods.
Now, the multivariate calibration equation can be written in a
matrix form as
(18)
By solving simultaneously the set of equations described by
(18), the unknown polynomial coefficients vector can be de-termined. Table III shows the obtained values of the polynomial
coefficients in (12).
Therefore, using (12), the moisture content of any unknown
rubber latex sample can be obtained. Fig. 14 shows the residues
(in standard deviations) of the three approaches described
above.
The standard error for the average method was 0.65%, for the
weighted average method was 0.49% and for the multivariate
regression method was 0.50%. It can be observed that despite
its increase complexity, multivariate regression model does not
produce better results. This is also visible in Fig. 14, where the
residues corresponding to the three methods show similar be-havior. For this reason, the present instrument implements the
weighted average method.
IV. CONCLUSION
In this paper, a dual-frequency microwave sensor based on
circular microstrip antenna has been described. Although the
sensor was tested on Hevea rubber latex solutions, tests on other
lossy dielectric liquids such as Soya milk, water-based paints,
and coconut milk are currently under study.
Three calibration schemes are described and tested. The
moisture content of the rubber latex sample could be de-
termined with a standard error of 0.49%. The sensor can
be used also for temperature independent moisture contentmeasurements.
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REFERENCES
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Wiley, 1971, pp. 764767.[4] F. L. Warner, Microwave Attenuation Measurement (IEEE Mono-
graph Series No 19). Stevenage, Herts: Peter Peregrinus, 1977, pp.272277.
[5] I. J.Bahland P.Bhartia, Microstrip Antennas. Norwood, MA: ArtechHouse, 1980, pp. 8591.
[6] N. Kumprasert and W. Kiranon, Simple and accurate formula for res-onant frequency of the circular microstrip disk antenna, IEEE Trans.
Antenna Propagat., vol. 43, no. 11, pp. 13311332, Nov. 1995.[7] B. K. Khalid, The application of microstrip sensors for determination
of moisture content in rubber latex, J. Microw. Power Electromagn.Energy, vol. 23, no. 1, pp. 4552, 1988.
[8] F. Daschner and R. Knochel, 2003 A new Transmission-Line Sensorfor Measuring the Composition of Foodstuffs Using Microwave, inProc. 5th Int. Conf. Electromagn. Wave Interaction with Water and
Moist Substances, Rotorua, New Zealand, Mar. 2326, 2003, pp. 524.
Mohamed Mustafa Ghretli was born in Tripoli,
Libya, in 1965. He received the B.S.E. (Hon) degreein applied physics from McMaster University,Hamilton, ON, Canada, in 1984, the M.Sc. degree
in applied optics from Windsor University, Windsor,ON, in 1992, and the Ph.D. degree in microwavephysics from the UniversitiPutra Malaysia, Selangor,in 2005.
Currently, he is a Lecturer at the Higher Collegefor Electronic Technology, Tripoli, Libya. His re-search interests include microwave moisture sensors,
microwave dielectric properties of materials, and computational physics.
Kaida Khalid was born in Terengganu, Malaysia, in1952. He received the B.Sc. degree in physics fromthe National University of Malaysia, CITY LO-CATION, in 1976, the M.Sc. degree in solid-statephysics from the University of London, London,
U.K., in 1978, and the Ph.D. degree in electronicand electrical engineering from the University ofBirmingham, Birmingham, U.K., in 1986.
Currently, he is a Professor of Microwave Physicsand Techniques with theDepartment of Physics, Uni-versiti Putra Malaysia, Serdang. His research inter-
estsinclude microwave moisture sensors, microwave dielectric properties of ma-terials, and high-power microwave applications. Most of his work is primarilyengaged with the development of moisture sensors for hevea rubber latex andoil palm fruit.
Ionel Valeriu Grozescu was born in Prahova,Romania, in 1965. He received the B.Sc. degree inphysics and the M.Sc. degree in optics and laserstechnology from Bucharest University, Bucharest,Romania, in 1989 and 1990, respectivley, and thePh.D. degree in photothermal phenomena from theUnivesity Putra Malaysia, Selangor, Malaysia, in1994.
He was with the Institute of Atomic Physics,Romania in 1990 involved with R&D activitiesconcerning the laser technology and numerical
simulations of optical nonlinear effects. He is currently an Associate Professorwith the Department of Physics, University Putra Malaysia. His research
activities includes modeling of laser induced thermal phenomena, design, andconstruction of instruments for measurement of thermal properties of liquids,remote control, and automation systems.
Mohd. Hamami Sahri was born in Batu PahatJohor, Malaysia, in 1953. He received the Bachelorof Forestry Sciences degree from the the UniversitiPertanian Malaysia (UPM), now known as theUniversiti Putra Malaysia, Serdang, in 1978, theM.S. degree in wood products engineering from theState University of New York (SUNY), College ofEnvironmental Science and Forestry, Syracuse, in
1981, and the Ph.D. degree in wood anatomy fromUniversiti Kebangsaan Malaysia, Bangi Selangor, in1995.
He started his teaching career with UPM in 1981. He was appointed as Headof theDepartment of ForestProduction, UPM, from1996 to 1999, andpromotedto Associate Professor in 1997. He is currently the Professor of Wood Anatomyand Quality and the Dean of the Faculty of Forestry, UPM. His research inter-ests include characterization of forest plantation and less-used timber species ofMalaysia. Lately, he conducted more research works on lignocellulosic mate-rials from agricultural and forestry resources.
Zulkifly Abbas was born in Alur Setar, Malaysia,
in 1962. He received the B.Sc. degree (Hon) inphysics from the University of Malaysia, KualaLumpur, in 1986, the M.Sc. degree in microwaveinstrumentation from the Universiti Putra Malaysia(UPM), Serdang, in 1994, and the Ph.D. degree
in electronic and electrical engineering from theUniversity of Leeds, Leeds, U.K., in 2000.
Currently, he is a Senior Lecturer with the Depart-ment of Physics, UPM, where he has been a facultymember since 1987. He is also the Head of the Labo-
ratory of Mathematical Sciences and Applications, Institute for MathematicalResearch, UPM, where he is the Leader of the Wave Propagation ResearchGroup. His main personnel research interest is in the theory, simulation, andinstrumentation of electromagnetic wave propagation at microwave frequenciesfocusing on the developmentof microwavesensors for agricultural applications.