rf signal processing planar antennas for beam tracking and...

RF Signal Processing Planar Antennas

for Beam Tracking and

Direction-of-Arrival Estimation

(ビーム追尾および到来方向推定用 RF 信号処理平面アンテナに関する研究)

RIMI RASHID Student ID: 15634013

A Thesis submitted in partial fulfillment of the requirements for the degree of

Doctor of Philosophy in Electrical and Electronic Engineering To

Graduate School of Science and Engineering Saga University

Supervised by

ICHIHIKO TOYODA

Professor, Saga University

March, 2018

RF Signal Processing Planar Antennas

for Beam Tracking and

Direction-of-Arrival Estimation

(ビーム追尾および到来方向推定用 RF 信号処理平面アンテナに関する研究)

RIMI RASHID Student ID: 15634013

Graduate School of Science and Engineering

Saga University, Saga, Japan

ii

Abstract

RF Signal processing concerns the analysis, synthesis, and modification of radio

frequency signals. In this thesis, planar antenna arrays are combined with RF

signal processing technology which brings a new era to research field on planar

antennas. Microwave integrated circuits (MIC) i.e. magic-T, RF multiplier etc.

are integrated with same planar structure which makes the whole antenna

structure very simple and compact. Several antennas have been developed to

serve beam tracking operation and direction-of-arrival estimation purpose. The

antennas are designed and their basic concepts are confirmed by their

experimental results. The antennas designed in this research work employ phase

monopulse mechanism.

A planar beam tracking antenna is proposed and it consists of a magic-T, two

antenna elements and phase shifters. The basic concept of the antenna is to adjust

the phase shifter using the difference of the signals received by the two antenna

elements and shift the beam to the direction of arrival wave. The prototype of the

antenna is made and beam tracking function is confirmed.

DOA estimation antenna has been designed to enhance the estimation capability

by integrating RF multiplier. RF Multiplier detects the phase relation between

sum and difference of the two received signals. The proposed antenna provide

wide range of DOA estimation whereas conventional monopulse DOA

estimation antennas determine the angle of half space. The proposed concept is

verified by fabricating the prototype of this antenna.

Multilayer structure of DOA estimation antenna is also proposed in this

research work. The proposed antenna can detect DOA estimation in two plane

i.e. xz-plane and yz-plane. To obtain dual axis angle of arrival detection, a

multilayer structure is introduced. The proposed structure consists of four

annular slot antennas as antenna elements and magic-Ts. The prototype of this

antenna is fabricated and confirmed its perception.

iii

Acknowledgement

First and foremost, I would like to express my deepest gratitude to my adviser

Dr. Ichihiko Toyoda, Professor, Department of Electrical and Electronic

Engineering, Saga University, for accepting me as a Ph.D student. It was

impossible for me to complete my research without his encouragement,

thoughtful guidance and critical comments. Dr. Toyoda has supported me not

only by providing a research assistantship over almost three and half years,

but also encouraged me academically and emotionally through the rough road

to finish this thesis. He has been supportive since the day I began working

in this lab. It is also a great honor and privilege for me to work under his

supervision.

I would like to thank Dr. Eisuke Nishiyama, Associate Professor,

Department of Electrical and Electronic Engineering, Saga University, for

providing valuable guidance, technical supports and mentorship during this

research work. I also wish to pay my gratefulness to Dr. Takayuki Tanaka,

Associate Professor, Department of Electrical and Electronic Engineering,

Saga University, for his fruitful discussions, encouragement, supports and

motivational speech. I would also like to thank Mr. Tasuku Uechi for his

valuable support.

I am thankful to Dr. Shinichi Sasaki and Dr. Sumio Fukai, Associate

Professor, Department of Electrical and Electronic Engineering, Saga

University for their valuable comments and suggestions during my thesis

defense presentation.

iv

I would like to express my gratitude and thanks to all faculty members of

Electrical and Electronic Engineering Department and the students of my

laboratory and for their outstanding support during all these days at Saga

University. I would also like to show my gratitude to Dr. Hiroshi Satow, Dr.

Md. Azad Hossain and Muhammad Asad Rahman for their support and

suggestions to peruse my research activities.

I am especially indebted to the authority of Saga University and the

government of Japan for providing financial support through

Monbukogakusho Scholarship to carry out my research work.

I would also like to thank to my parents, my parents in-laws and younger

brother whose love and guidance are with me in whatever I pursue. They were

always supporting me and encouraging me with their best wishes.

Most importantly I would like to thank my husband Nazmush Shams. He

was always there cheering me up and stood by me through the good and bad

timing.

v

Contents Abstract ii

Acknowledgements iii

List of Tables viii

List of Figures ix

1 Introduction 1

1.1 Background 1

1.2 Research Approach 5

1.3 Organization of this Thesis 6

2 Microstrip Planar Antenna 8

2.1 Planar Antenna 8

2.2 Feeding Techniques 10

2.2.1 Microstrip Line Feeding 10

2.2.2 Coaxial Feeding 11

2.2.3 Aperture Coupling 11

2.2.4 Proximity Coupling 11

2.3 Planar Array Antenna using Both-Sided MIC Technology

12

2.3.1 Microstrip-slot Branch 14

2.3.2 Slot-Microstrip Branch 14

2.3.3 Magic-T 14

3 5.8 GHz Beam Tracking Antenna using Magic-T 18

vi

3.1 Introduction 18

3.2 Basic Principle 20

3.2.1 Theoretical Evaluation 23

3.3 Antenna Design 26

3.3.1 Phase Shifter 26

3.3.2 Magic-T 31

3.4 Prototype E-plane Beam Tracking Antenna 33

3.5 Result and Discussion 33

3.6 H-plane Beam Tracking Antenna Design 38

3.6.1 Operating Principle 38

3.6.2 Antenna Structure 40

3.6.3 Prototype Antenna 42

3.6.4 Result and Discussion 42

3.7 Summary 49

4 DOA Estimation Antenna 50

4.1 Introduction 50

4.2 Operating Principle 53

4.3 Antenna Structure 58

4.3.1 RF Multiplier 59

4.3.2 Power Divider 64

4.4 Measured Result 67


4.4.2 Result and Discussion 68

4.5 Summary 73

5 Dual Axis DOA Estimation Antenna 74

5.1 Introduction 74

5.2 Basic Operation Principle 77

5.3 Antenna Structure 78

5.4 Result and Discussion 79

5.4.1 Simulated Result 79


vii

5.4.3 Measured Result 82

5.5 Conclusion 84

6 Conclusion 85

References 87

Appendix 93

viii

List of Tables 3.1 Comparison between calculated and measured beam

directions 36

ix

List of Figures 1.1 Research Approach 7

2.1 Microstrip planar antenna 9

2.2 Microstrip line feed 9

2.3 Coaxial feed 9

2.4 Aperture coupling feed 9

2.5 Proximity coupling feed. 9

2.6 Microstrip-slot Branch Circuit 13

2.7 Slot-Microstrip Branch Circuit 13

2.8 Basic Structure of Magic-T 15

2.9 Equivalent Circuit of Magic-T 15

2.10 Simulated performance of the magic-T 17

3.1 Block diagram of the proposed beam tracking concept. A magic-T generates the sum () and difference (Δ) of the signals received by two antennas. The Δ signal is used to control the phase shifter.

21

3.2 Operating principle of the proposed beam tracking antenna. 22

3.3 Vector diagram of the proposed beam tracking antenna. 22

3.4 Theoretical radiation pattern of . The main beam of the shift to right according to the phase shift value 0 . The

radiation pattern of a single rectangular patch antenna is D also plotted as a reference.

24

3.5 Theoretical radiation pattern Δ signals. The null of the Δ shift to right according to the phase shift value 0 . The

24

x

radiation pattern of a single rectangular patch antenna is D also plotted as a reference.

3.6 Structure of the proposed beam tracking antenna. A magic-T and two phase shifters employing the both-sided MIC technology are integrated with two antenna elements. The bias voltage of each phase shifter is applied via a bias probe at the center of the antenna element.

25

3.7 Phase shift vs capacitance of the varactor diode. 27

3.8 Structure of the phase shifter. The phase shifter consists of a de Ronde’s coupler and varactor diodes.

28

3.9 Prototype of the 5.8-GHz phase shifter. 28

3.10 Measured reflection coefficient of the prototype phase shifter.

29

3.11 Measured transmission coefficient of the prototype phase shifter.

29

3.12 Phase shift vs applied bias voltage relation. 30

3.13 Schematic structure of the magic-T. 31

3.14 Prototype of the 5.8-GHz beam tracking antenna (size: 198 × 131 mm).

32

3.15 Experimental setup at anechoic chamber. 33

3.16 Measured return loss at Port 1. 34

3.17 Measured return loss at Port 2. 34

3.18 Measured gain at Port 1. 35

3.19 Measured gain at Port 2. 35

3.20 Measured gain with respect to the beam angle. 37

3.21 Comparison of the beam shift characteristics of 5.8- and 10-GHz antennas.

37

3.22 Configuration of magic-T integrated antennas. 39

3.23 Antenna structure of the H-plane beam tracking antenna. 40

3.24 Prototype of the 5.8-GHz H-plane beam tracking antenna (size: 110 110 mm

41

3.25 Reflection coefficient at Port 1 (). 43

3.26 Reflection coefficient at Port 2 (). 44

3.27 Radiation pattern of signal. 45

3.28 Radiation pattern of signal. 46

3.29 Beam angle vs applied voltage plot. 47

xi

3.30 Gain vs beam angle plot. 47

4.1 Block Diagram of the proposed DOA estimation antenna concept. Sum () and diference () of the two received signals are generated by a magic-T. Sign of signal can be obtained from an RF multiplier.

54

4.2 Basic operating principle of the proposed DOA estimation antenna.

55

4.3 Structure of the proposed planar DOA estimation antenna. Two magic-Ts and an RF multiplier employing the both-sided MIC technology are integrated with four antenna elements.

58

4.4 Structure of the RF multiplier. Sum () and difference () of the two received signals are fed to Port RF1 and RF2 respectively. By observing the output voltage, sign of signal can be perceived.

59

4.5 Two diode balanced multiplier. 60

4.6 Theoretical relation between the RF multiplier output voltage and arrival angle

63

4.7 Two input two output power dividers used to divide the powers to RF ports and RF multiplier.

64

4.8 Two input two output power dividers used to divide the powers to RF ports and RF multiplier.

66

4.9 Prototype 10-GHz DOA estimation antenna (Size: 113113 mm, f =10 GHz).

67

4.10 Measured reflection coefficient plots of Port 1 and Port 2 of the designed antenna.

69

4.11 Simulated and measured radiation patterns of and Δ signals.

69

4.12 Relation between / and arrival angle . Arrival angle can be determined from this plot and sign of the signal.

70

4.13 Experimental Orientation in Anechoic Chamber to measure output of RF multiplier.

70

4.14 Measured RF multiplier output voltage vs. arrival angle (Frequency Dependence).

72

4.15 Measured RF multiplier output voltage vs. arrival angle (Power density dependence).

72

5.1 Block operating principle of the DOA estimation Antenna concept.

76

xii

5.2 Concept of the dual-axis DOA estimation antenna. 76

5.3 Schematic structure of the antenna (5.8 GHz). 78

5.4 Simulated reflection coefficient plot of Port 1, 2, 3 and 4. 80

5.5 Simulated radiation pattern plot of and signal in xz-plane and yz-plane.

80

5.6 Prototype of the 5.8-GHz dual axis DOA estimation antenna. (size: 88mm × 88mm)

81

5.7 Experimental setup in anechoic chamber. 83

5.8 Measured reflection coefficient plot of Port 1, 2, 3 and 4. 83

5.9 Measured radiation pattern plot of and signal in xz-plane and yz-plane.

84

1

Chapter 1

Introduction

1.1 Background

Wireless communication is a transmission of information over a distance

without requiring wires, cables or any other electrical conductors. Wireless

communication is one of the important mediums of transmission of data or

information to other devices. Antennas are key components of any wireless

communication system [1,2,3]. The IEEE standard definitions of terms for

antenna defines the antenna as “a means for radiating or receiving waves” [4].

The principles necessary for antennas were originated by Maxwell [3] in

1864 and based on these principles Hertz’s first antenna was developed in

1887 [5]. He performed transmit receive experiments using dipoles, loops and

reflector antennas. After that in 1901, Marconi succeeded in sending signals

over large distance from England to Newfoundland. Until 1940s antenna

technology was primarily focused on wire related radiation elements. After

World War II, modern antenna technology was born and new elements such

as waveguide aperture, horns, reflectors, lenses etc. were first introduced [6]

The concept of microstrip patch antenna originated in the 1953s [7], but it

did not attract serious attention until the 1970s. The microstrip patch antenna

was introduced by Munson in 1972 [9]. In 1974, his work was published in a

journal paper. His work discussed about the wrap around microstrip antenna

and the rectangular patch [10]. Shortly after Munson’s work, Howell also

2

discussed rectangular patch antenna and after that he introduced the circular

patch as well as the circularly polarized patch antenna [11]. Soon after the

introduction of microstrip antenna, different methods to analyze this antennas

including the transmission line model [12], the cavity model [13] and the

spectral domain method [14] were reported in different literatures.

Microstrip antennas are widely used in different microwave

communication applications because of their simplicity and compatibility

with printed circuit technology. Microstrip antennas usually have the

important advantages of being low profile and conformable. The main

disadvantage of microstrip antenna compared with other antennas are lower

radiation efficiency and small bandwidth, though many specialized

techniques have been developed to increase the radiation efficiency and

bandwidth of microstrip antenna [3].

A wide range of wireless technologies is currently used in wireless

communication systems. Advanced technologies based on digital signal

processing have developed to fulfill the requirements of higher data rate,

larger capacity and higher efficiency. Accordingly, multifunction and high

performance microwave and millimeter wave components are also required

to promote the wireless technologies [8]. Development of advanced planar

antennas based on RF signal processing is a revolution in the field of the

wireless communication technology. These new generation antennas provide

many important features such as low profile, light weight, low cost and ease

of integration into arrays. These features make them ideal components for

radars and modern communication systems.

3

Radar is an object-detection system which utilizes electromagnetic waves,

specifically radio waves. At the beginning, main function of a radar was to

detect object from long distances. After that, along with the development of

the radar technology, the radar has multiple functions such as simultaneous

search and tracking of multiple targets, and collects various information

related to the targets. It can determine the range, altitude, direction, or speed

of both moving and fixed objects such as aircraft, ships, spacecraft, guided

missiles, motor vehicles and weather formations.

An automotive radar is used to locate objects in the vicinity of the car and

consists of a transmitter and a receiver. The transmitter sends out radio waves

that hit an object and bounce back to the receiver. By controlling the

direction in which radio waves are sent and received, it is possible to detect

objects' distance, speed and direction. This requires steerable antennas that

can be automatically directed or received antennas so that signals receive

simultaneously from several different directions. These devices are used in

advanced cruise control systems, collision warning systems, blindspot

monitoring, lane-change assistance, rear cross-traffic alerts, and back-up

parking assistance. More recently, advancements in radar technology have

allowed these systems to have the functionality of more preventative safety

features such as collision mitigation. The main functions of automotive radar

system are range estimation, doppler frequency estimation, direction of

arrival (DOA) estimation and tracking.

One of today’s major challenges is to develop such antennas for radars with

high enough performance at a reasonable cost. To accomplish the above

4

requirements, advanced planar antenna technology is fully utilized and novel

planar antennas are developed for beam tracking and direction of arrival

estimation purpose. In this thesis, beam tracking antennas and direction of

arrival estimation antennas are described.

DOA estimation antenna can be used to detect the angle of the signal to

track the position of the object. There are several types of angle measurement

techniques, such as beam switching method and phased array method. In the

beam switching method, angle measurement is performed by switching fixed

beams. In the phased array system, a number of antennas are used to change

the shape of the beam. These systems become large sized and not suitable for

mounting on a moving body. There is another monopulse method for angle

measurement method. In the monopulse method, one signal is received by

two antennas, and the angles are measured using the sum and difference

values (Σ and Δ). Since the monopulse method is an angle measurement

method that can easily be used compared to other two principles, the

monopulse method is focused in this research. The beam tracking antennas

presented in this thesis is also based on the monopulse mechanism.

Many research activities have been presented on monopulse antennas in

recent years. A single layer monopulse microstrip antenna array was proposed

to achieve two dimensional monopulse performance [15]. Four 3-dB hybrid

couplers and several 90 delay lines are used as a comparator in this design.

A multilayer 44 microstrip patch antenna has been demonstrated to extend

the impedance bandwidth of the antenna in C-band [16]. Another monopulse

antenna has been designed by integrating an electromagnetically coupled

5

(EMCP) MSA array structure and a rat-race hybrid instead of a branch line

coupler as a comparator so that it can be used as a transmitter in addition to a

receiver [17]. A dual probe fed single aperture monopulse antenna has been

presented to eliminate complex phasing networks and aperture arrangement

of the antenna. This structure consists of a slot embedded patch, two probe

inputs and a 180 directional coupler [18]. The DOA estimation concept based

on monopulse mechanism has been already discussed in [19]. The classical

monopulse concepts are extended to a general complex monopulse concept

which can utilize information from side lobes by using phase shift of the

signals.

There are several problems with the arrival angle estimation and tracking

devices. First of all, it is size and weight. Large or heavy items are not suitable

for moving objects. Therefore, the devices should be small in size. Besides,

many tracking and arrival angle estimation devices are complex and

expensive. In order to realize it at low cost, simplification of the structure

becomes a problem. In this research, the aim is to realize beam tracking and

arrival angle estimation configuration with a compact, high performance and

simple configuration by using both plane circuit technology [20].

1.2 Research Approach

To realize the increasing demand of RADAR detection and tracking

techniques, low cost planar antennas are proposed in this thesis. Microwave

integrated circuits technology is applied with antenna array structure. Figure

1.1 illustrates the basic structure of the research work presented in this thesis.

6

Two beam tracking antennas consisting of magic-T and phase shifter have

been designed. Varactor diode loaded phase shifter is used to track the beam

by changing the capacitance of the diode. On the other hand, a 2×2 slot

antenna array with orthogonal feed networks combining with magic-T has

been implemented for DOA estimation at both planes. Another DOA

estimation antenna that shows the extended angle performance has also been

designed using RF mixer.

1.3 Organization of this Thesis

In this dissertation, chapter 1 contains the background and objective of the

proposed research. The introduction and fundamental of the microstrip planar

antenna is described in chapter 2. Microwave integrated circuits i.e. magic-T

and de Ronde’s coupler are also described in this chapter. Chapter 3 describe

the concept and outcomes of the beam tracking antenna. Both E-plane and H-

plane beam tracking operation are described two separate antenna structures.

7

Chapter 4 describes the basic principle and realization of the extended angle

of DOA estimation antenna. The proposed DOA estimation antenna can

measure the arrival angle broader than conventional DOA estimation antenna.

Chapter 5 discuss about the dual axis DOA estimation antenna. Arrival angle

can be estimated in xz-plane and yz-plane by this antenna. Finally, in chapter

8, the conclusion, findings and scope of future work are narrated.

Figure 1.1 Research Approach

Antenna Array

Phase Shifter RF Mixer

Magic-T

Beam Tracking Antenna

Dual-axis DOA

Estimation Antenna

Extended Angle DOA Estimation

Antenna

8

Chapter 2

Microstrip Planar Antenna

2.1 Planar Antenna

Microstrip antennas have been one of the most innovative topics in antenna

theory and designs in recent years, and are increasingly finding application in

a wide range of modern microwave systems. As shown in Figure 2.1, the basic

configuration of a microstrip antenna is a metallic patch printed on a thin,

grounded dielectric substrate. Originally, the element is fed with either a

coaxial line through the bottom of the substrate, or by a coplanar microstrip

line. This type of excitation allows feed networks and other circuitry to be

fabricated on the same substrate as the antenna element. The microstrip

antenna radiates a relatively broad beam broadside to the plane of the

substrate. Thus the microstrip antenna has a very low profile, and can be

fabricated using printed circuit (photolithographic) techniques. This implies

that the antenna can be made conformable, and potentially at low cost. Other

advantages include easy fabrication into linear or planar arrays, and easy

integration with microwave integrated circuits.

To a large extent, the development of microstrip antennas has been driven

by system requirements for antennas with low-profile, low-weight, low-cost,

easy integrability into arrays or with microwave integrated circuits, or

polarization diversity. Thus microstrip antennas have found application in

both the military and the civil sectors.

9

Figure 2.1 Microstrip planar antenna.

Figure 2.2 Microstrip line feed.

Figure 2.3 Coaxial feed.

Figure 2.4 Aperture coupling feed.

Figure 2.5 Proximity coupling feed.

Patch

DielectricSubstrate

Ground

Patch

Microstrip line feed

Dielectric Substrate

Patch

GroundCoaxial

Connector

Patch

DielectricSubstrate

Ground

Slot Line

Microstrip Feed Line

Patch

DielectricSubstrate

Microstrip Feed Line

10

Disadvantages of the original microstrip antenna configurations include

narrow bandwidth, spurious feed radiation, poor polarization purity, limited

power capacity, and tolerance problems. Much of the development work in

microstrip antennas has thus gone into trying to overcome these problems, in

order to satisfy increasingly stringent systems requirements. This effort has

involved the development of novel microstrip antenna configurations, and the

development of accurate and versatile analytical models for the understanding

of the inherent limitations of microstrip antennas, as well as for their design

and optimization.

2.2 Feeding Techniques

There are many different methods of feeding and four most popular methods

are microstrip line feed [21], coaxial probe [22], aperture coupling [23] and

proximity coupling [24].

2.2.1 Microstrip Line Feeding

As shown in Figure 2.2, microstrip line feed is one of the easier methods to

fabricate as it is a just conducting strip connecting to the patch and therefore

can be consider as extension of patch. It is simple to model and easy to match

by controlling the inset position. However the disadvantage of this method is

that as substrate thickness increases, surface wave and spurious feed radiation

increases which limit the bandwidth.

11

2.2.2 Coaxial Feeding

Coaxial feeding, illustrated in Figure 2.3, is a feeding method in which that

the inner conductor of the coaxial is attached to the radiation patch of the

antenna while the outer conductor is connected to the ground plane.

Advantages of this method are easy of fabrication, easy to match, and low

spurious radiation. Narrow bandwidth is the main disadvantage of this

method. Moreover, it is difficult to model specially for thick substrate and

possess inherent asymmetries which generate higher order modes which

produce cross-polarization radiation.

2.2.3 Aperture Coupling

Aperture coupling shown in Figure 2.4 consist of two different substrate

separated by a ground plane. On the bottom side of lower substrate there is a

microstrip feed line whose energy is coupled to the patch through a slot on

the ground plane separating two substrates. This arrangement allows

independent optimization of the feed mechanism and the radiating element.

Normally top substrate uses a thick low dielectric constant substrate while for

the bottom substrate; it is the high dielectric substrate. The ground plane,

which is in the middle, isolates the feed from radiation element and minimizes

interference of spurious radiation for pattern formation and polarization purity.

It allows independent optimization of feed mechanism element.

2.2.4 Proximity Coupling

Proximity coupling shown in Figure 2.5 has the largest bandwidth, has low

spurious radiation. However fabrication is difficult. Length of feeding stub

12

and width-to-length ratio of patch is used to control the match.

2.3 Planar Array Antenna Using Both-Sided MIC

Technology

Design of the feed network is one of the most important part of the planar

array technology. Generally, microstrip feed lines are used to design

conventional technique to design feed network. In case of microstrip feed

network, most of the feed circuits are parallel. This increases the feed loss

with the increase of the size of the array antenna. Besides, many matching

circuits are required, as all the feeding circuits are connected in parallel. In

the result, the feeding circuit layout for the array antennas becomes

complicated and the length of the feed lines becomes much longer. Those

cause the array antenna to increase the feeding loss as well as an undesired

radiation. To overcome these problems, the Both-Sided MIC technology [20]

is a good candidate for designing the array antenna feed. The array antenna

needs no impedance matching circuits and has a very simple circuit

configuration mainly due to the excellent combination of both the microstrip-

slot parallel branch circuit and the slot-microstrip series branch circuit.

13

(a) Configuration

(b) Equivalent circuit

Figure 2.6 Microstrip-slot branch.

(a) Configuration

(b) Equivalent circuit

Figure 2.7 Slot-microstrip branch.

Microstrip Line

Port 1Z1

Port 3Z3

Port 2Z2

Slot Line

Port 2Z2

Port 3Z3

Port 1Z1

Slot Line

Microstrip Line

Microstrip Line

Slot Line

Port 1Z1

Port 2Z2

Port 3Z3

Port 3Z3

Port 2Z2

Port 1Z1

Microstrip Line

Slot Line

14

2.3.1 Microstrip-slot Branch

Figure 2.6 shows the schematic layout and equivalent circuit for a microstrip-

slot branch. The circuit is composed of a microstrip line on a substrate and a

slot line on the ground plane. The impedance for port 1, port 2 and port 3 are

considered as Z1, Z2 and Z3, respectively. The microstrip-slot branch are

connected in parallel, so the condition to match impedances is Z2 = Z3 = 2Z1.

In addition, two output signals at the equal distance points from the branch

point on the slot line are same amplitude and in phase.

2.3.2 Slot-Microstrip Branch

Figure 2.7 shows the schematic diagram and equivalent circuit for a slot-

microstrip branch. The impedance for port 1, port 2 and port 3 are considered

as Z1, Z2 and Z3, respectively. The slot-microstrip branch circuits are

connected in series, so the condition for impedance matching is Z2 = Z3 =

Z1/2. Thus, at the equal distance points from the branch point in the slot line,

two output signals are same amplitude and anti-phase.

2.3.3 Magic-T

Integrated magic-T structures have been widely used at

microwave/millimeter wave frequencies as functional components of

complex microwave circuits and systems.The magic-Ts are four-port devices

that offer in-phase and anti-phase signal division between their two output

ports [25-31]. The main applications of magic-T are balanced-mixers,

discriminators, interferometers, and beam-forming networks. Some desired

15

properties of magic-Ts include wide bandwidth phase and amplitude balance,

low insertion loss, high isolation, compact size, and simplicity of its

fabrication.

Figure 2.8 shows the basic structure of the magic-T. It is the combination

of a microstrip line T junction and a slot-to-microstrip line branch. Figure 2.9

shows the equivalent circuit of the magic-T. The impedance of Port M1 is half

of the impedance of Port M3 and M4. On the other hand, the impedance of

Port M2 is double of the impedance of Port M3 and M4. The signals fed from

Figure 2.8 Basic Structure of Magic-T

Figure 2.9 Equivalent Circuit of Magic-T

Microstrip Line

Microstrip Line T Junction

Microstrip Line

Slot Line

Slot to Microstrip Line Branch

Magic-T

Microstrip Line

Slot Line

Port M3 (Z0)

Port M1(Z0/2)

Port M4 (Z0)

Port M2 (2Z0)

Anti-Phase Signal

In-Phase Signal

Port M1(Z0/2)

Microstrip Line

Port M3 (Z0)

Port M4 (Z0)

Port M2 (2Z0)

Slot Line

16

Port M3 and M4 with the same phase are combined and output to Port M1, as

the microstrip line T junction is a parallel branch. On the other hand, as the

slot-to-microstrip line branch is a series branch, the signals fed from Port M3

and M4 with the anti-phase are combined and output to port M2. Furthermore,

isolation between Port M1 and M2 are achieved due to the difference of the

propagation modes of the microstrip line and slot line.

Figure 2.10 show the simulated performance of the magic-T. As shown in

Figure 2.10(a), the return losses of Port M1 and Port M2 are better than 20

dB at 5.8 GHz, while the insertion losses of the in-phase signal ( 31S , 41S ) and

anti-phase signal ( 32S , 42S ) are less than 3.1 dB and 4.1 dB, respectively. The

isolations ( 34S , 12S ) are better than -19.9 dB and -84.1 dB. Extremely high

isolation between Port M1 and M2 is achieved. Figure 2.10(b) shows the

phase difference vs. frequency plot, which confirmed that the magic-T

provides in-phase and anti-phase power division. Then, when RF signals are

fed from Port M3 and M4, the sum and difference of the signals are separately

obtained at Port M1 and M2, respectively. The simulation is performed by

Keysight Technologies’ ADS simulation software.

17

(a) Amplitude of S-parameters

(b) Phase difference

Figure 2.10 Simulated performance of the magic-T

-40

-30

-20

-10

0

S-p

aram

eter

s [d

B]

7.06.56.05.55.04.5Frequency [GHz]

S11 S22

S31, S41

S32, S42

270

180

90

0

-90

Pha

se D

iffe

renc

e [d

eg.]

7.06.56.05.55.04.5Frequency [GHz]

S31-S41

S32-S42

18

Chapter 3

5.8 GHz Beam Tracking Antenna Using

Magic-T

This chapter discusses a novel planar beam tracking antenna and brings a new

prototype antenna in wireless communication systems. The proposed antenna

consists of a magic-T, two antenna elements and two phase shifters. The main

idea for the antenna is to adjust the phase shifter using the difference of the

signals received by the two antenna elements to tilt the beam in the direction

of the arrival wave. Theoretical discussion is presented to explain the concept.

Both-sided MIC technology is effectively used to integrate the magic-T and

the phase shifters with the antenna elements in a simple structure. A prototype

antenna of the new design is fabricated and the radiation pattern and return

loss are measured. Simulation and experimental results of the beam direction

vs. applied voltage are successfully compared and the proposed concept is

experimentally demonstrated.

3.1 Introduction

Development of advanced planar antennas based on RF signal processing

is a revolution in the field of the wireless communication technology. These

new generation antennas provide many important features such as low profile,

19

light weight, low cost, and ease of integration into arrays. These features

make them ideal components for radars and modern communication systems,

specially, for the portable wireless devices. Properties of planar antennas have

been discussed many times in many journals and proceedings [32, 33].

Several types of advanced planar antennas have been also developed for the

RF signal processing [35]. These antennas are constructed with microstrip

lines and slot lines on both sides of the substrate and microwave circuits such

as a magic-T are integrated with antenna elements [20]. A direction of arrival

estimation antenna [36-38] and beam steering antenna [39] are already

presented.

Several research works have done to enhance the evolution of beam

tracking systems in wireless communication. There are many tracking

systems i.e. manual/programmed steering, monopulse tracking, sequential

amplitude sensing tracking (conical scan and step track) and electronic beam

squinting are developed. A planar antenna for a beam tracking system, was

proposed for land vehicle satellite communications. A step back method by

using an angular rate sensor is used in this system [40]. Another tracking

phased array antenna system for the shipboard station in X-band satellite

communication was described. This antenna has TX and RX antenna beams

as well as tracking beam where beams are independently steered

electronically in elevation and mechanically in azimuth [41].

We have also presented a planar beam tracking antenna using a magic-T

applying monopulse tracking mechanism [42]. The main function of this

beam tracking antennas is shifting their beam according to the received waves.

20

The beam tracking antenna we have proposed is also a novel antenna in the

RF signal processing. The antenna was designed, fabricated and also

measured experimentally.

In this chapter, a novel 5.8-GHz planar beam tracking antenna is proposed.

The advantage of the proposed antenna over digital processing beam tracking

is low power consumption and rapid response due to the analog processing of

the system. The concept and basic operation principle of the beam tracking

antenna are discussed with theoretical calculations in section 3.2. In section

3.3, the structure and design of the antenna which integrates antenna elements

and microwave circuits such as a magic-T and phase shifter are described.

These microwave circuits are also briefly introduced in this section. Section

3.4 shows a prototype beam tracking antenna designed for 5.8-GHz band. The

measured results which demonstrate the concept of the proposed beam

tracking antenna and the comparison between simulation and experimental

results are highlighted in this section 3.5. Section 3.6 describe the basic

concept, structure and measured result of the H-plane beam tracking antenna.

3.2 Basic Principle

Figure 3.1 illustrates the basic block diagram of the proposed beam tracking

concept. The proposed beam tracking antenna consists of two antennas, a

magic-T and a phase shifter. A detector and controller circuit are attached to

the proposed antenna to adjust the phase shifter. The phase shifter is used to

change the phase of the signal received by one of the two antenna elements.

By changing the phase shift value of the phase shifter to compensate the phase

21

difference, maximum power is obtained by combining the two received

signals. It corresponds that the beam of the antenna array shifts in the

direction of the arrival wave. In this proposed antenna, a magic-T is employed

to combine the two received signals and detect the phase difference of the

signals received by the two antennas. As a magic-T is a microwave circuit

that provides in-phase or anti-phase power division according to the input and

output ports, the sum () and difference (Δ) of the received signals are easily

obtained by using the magic-T.

Figures 3.2 and 3.3 describe the operating principle of the proposed

antenna. When arrival waves are received by the two antennas with angle

as shown in Figure 3.2, the waves have a phase difference φ. The relation of

Figure 3.1 Block diagram of the proposed beam tracking concept. A

magic-T generates the sum () and difference (Δ) of the signals received

by two antennas. The Δ signal is used to control the phase shifter.

Arrival wave

Detector & Controller

Circuit

Receiver

∑ ∆

Magic-T

Phase Shifter0

22

the arrival angle and the phase difference φ is expressed by the following

equation from phase monopulse mechanism concept.

sin

2 d (3.1)

where d and are the antenna separation and wavelength, respectively. Hence,

the sum () and difference (Δ) of the received signals are expressed in the

following expressions:

Figure 3.2 Operating principle of the proposed beam tracking

antenna [Reproduced courtesy of The Electromagnetics Academy].

Figure 3.3 Vector diagram of the proposed beam tracking antenna

[Reproduced courtesy of The Electromagnetics Academy].

12

1d 2

d sin

Arrival wave

#1

#2

Re

Im

23

2cose)(2

e)(ee)(

0

0

0

2

22

j

jjj

θD

θDθD (3.2)

2sine)(2

e)(ee)(

0

0

0

2

22

j

jjj

θjD

θDθD (3.3)

where φ0 and D() are the phase shift value of the phase shifter and the

radiation pattern of a single antenna element, respectively.

The main objective of this antenna is to shift the main beam direction

according to the arrival wave, i.e., make the maximum. Figure 3.3 shows

the relation among two received signals, their sum () and difference (Δ) with

a vector diagram. As shown in this figure and above expressions, becomes

maximum and Δ becomes zero when φ0 = φ. This means that the beam tilts in

the direction of the arrival wave by adjusting the phase shifter to make the Δ

minimum.

3.2.1 Theoretical Evaluation

Figures 3.4 and 3.5 show the theoretical results of the radiation pattern of the

and Δ signals calculated using Eqs. (3.2) and (3.3) for the phase shift φ0 =

0°, 30°, 60° and 90°. D() is also plotted as a reference. Here, the radiation

pattern of a single rectangular patch antenna is used for D(), which is

expressed by the following expression [43]:

24

θβL

θD sin2

cos)( (3.4)

Figure 3.4 Theoretical radiation pattern of . The main beam of the shift to right according to the phase shift value 0 . The radiation pattern of a

single rectangular patch antenna is D also plotted as a reference


Figure 3.5 Theoretical radiation pattern Δ signals. The null of the Δ shift to right according to the phase shift value 0 . The radiation pattern of a

single rectangular patch antenna is D also plotted as a reference


-40

-30

-20

-10

0

Rel

ativ

e P

owe

r [d

B]

-90 -60 -30 0 30 60 90

Angle[deg.]

D(= 0

o

= 30o

= 60o

= 90o

-40

-30

-20

-10

0

Rel

ativ

e P

owe

r [d

B]

-90 -60 -30 0 30 60 90

Angle[deg.]

D(= 0

o

= 30o

= 60o

= 90o

25

where, and L are the free space phase constant and patch length of the

microstrip antenna, respectively.

As the gradual increase of the phase shift φ0, the beam tilts to the right as

shown in Figure 3.4. The beam directions shifts to right from 0° to 6°, 12°

and 18° due to the increase of the phase shift value to 30°, 60° and 90°,

respectively. Similarly, null of the Δ is also shifted to the right as shown in

Figure 3.5.

Figure 3.6 Structure of the proposed beam tracking antenna. A magic-T

and two phase shifters employing the both-sided MIC technology are

integrated with two antenna elements. The bias voltage of each phase

shifter is applied via a bias probe at the center of the antenna element


Phase Shifter

A A’

Port 1

Microstrip Line

Slot Line

Port 2

Bias Probe

Magic-T

Σ

Δ

Varactor Diode

Capacitor

A A’

x

x

y

z

y

z

Patch

Impedance Transformer

26

3.3 Antenna Design

Figure 3.6 illustrates the structure of the proposed beam tracking antenna.

Two microstrip antennas, a magic-T and two phase shifters are integrated on

a substrate. Capacitors (9 pF) are used for DC cut. In this design, two phase

shifters are used to tilt the beam to both directions. The magic-T and phase

shifters are effectively employing the both-sided MIC technology [20]. The

input impedance of each antenna element is designed to be 100 Ω and it is

converted to the port impedances of 50 Ω using the magic-T and a quarter-

wavelength impedance transformer. Bias voltage of each phase shifter is

applied via a probe at the center of the patch to change the phase shift value.

3.3.1 Phase Shifter

In this proposed beam tracking antenna, beam can be shifted by adjusting

the phase shift value of the phase shifter. Phase shifter plays an important role

in this antenna. Phase shifter can be designed by integrating two varactor

diodes on the two coupled ports of a 90 hybrid coupler. A branch line coupler

(quadrature 90° hybrid) is a four-port network device with a 90° phase

difference between two coupled ports. The varactor diode acts like a variable

capacitor under reverse bias. By changing the capacitance of the varactor

diode the phase between the input and output port of the 90 hybrid coupler

can be changed.

27

Figure 3.7 shows the simulated performance of the phase shifter in different

frequencies. Phase shift value changes by varying the capacitance of the

varactor diodes. Simulation is done by using Keysight ADS simulation

software. During simulation process varactor diodes are replaced by capacitor.

Phase shift values are observed by varying the capacitance. Figure 3.7 implies

that phase shift value is zero at 2 pF capacitance. Phase shift value can

increase by decreasing the capacitor value. Phase shift vs capacitance

characteristics is same for any other frequency ranges i.e. 2.45 GHz, 5.8 GHz,

7.5 GHz and 10 GHz.

Figure 3.8 shows the structure of the phase shifter used in proposed 5.8-

GHz beam tracking antenna. In this configuration of the phase shifter,

varactor diodes are connected to the de Ronde’s coupler. The de Ronde’s

coupler is constructed with a microstrip line and slot line and it provides a 90

hybrid function with a simple structure. Two varactor diodes are attached to

Port R2 and R4 of the coupler and the phase shift is obtained by changing the

Figure 3.7 Phase shift vs capacitance of the varactor diode.

120

100

80

60

40

20

0

Pha

se S

hift

[Deg

.]

2.01.81.61.41.21.00.80.60.40.20.0

Capacitance [pF]

2.45-GHz 5.8-GHz 7.5-GHz 10-GHz

28

bias voltage of the varactor diodes. The signal received from the antenna

element is applied to Port R1 as the input and the output of the phase shifter

are received from Port R3 in different phase by applying different bias

voltages. Hence, the beam direction changes by changing the bias voltage.

Figure 3.9 shows the prototype of the 5.8-GHz phase shifter. Impedance

Figure 3.8 Structure of the phase shifter. The phase shifter consists of a de

Ronde’s coupler and varactor diodes.

Figure 3.9 Prototype of the 5.8-GHz phase shifter.

Port R1

Port R4

Port R2

Port R3

Microstrip Line

Slot Line

Varactor Diode

A A’

A’A

29

transformers are integrated to Port R1 and R3 for impedance matching

purpose. The bias voltage is applied to the varactor diodes via a bias-T. The

experiment procedure is done by using a network analyzer.

Figures 3.10 and 3.11 show the measured performance of the phase shifter.

The reflection coefficients of the phase shifter are below 10-dB level at 5.8

Figure 3.10 Measured reflection coefficient of the prototype phase shifter


Figure 3.11 Measured transmission coefficient of the prototype phase

shifter [Reproduced courtesy of The Electromagnetics Academy].

-30

-20

-10

0R

efle

ctio

n C

oeffi

cien

t [dB

]

7.06.56.05.55.04.5Frequency [GHz]

S11: 1 V 5 V 9 V S22: 1 V 5 V 9 V

-20

-15

-10

-5

0

Tra

nsm

issi

on C

oeffi

cien

t [dB

]

7.06.56.05.55.04.5Frequency [GHz]

S21: 1 V 5 V 9 V

30

GHz for every bias voltage as shown in Figure 3.10. It is also confirmed from

Figure 3.11 that the insertion losses are near 2 dB at the design frequency 5.8

GHz. Both results ensure the performance of the designed phase shifter very

well.

Figure 3.12 shows the relation between the phase shift value φ0 and applied

voltage of a prototype phase shifter. The red line indicates the simulation

result and blue line shows the measured result. In the simulation process,

relation between the phase shift value φ0 and applied voltage is calculated by

Keysight Technologies’ ADS simulation software. For this simulation

purpose, capacitors are used instead of the varactor diodes. The value of the

capacitor is tuned from 2 pF to 0.225 pF which is equivalent to 1 V to 10 V

(calculated by the data sheet of the varactor diode). Thus the phase shift value

φ0 can be obtained with respect to its applied bias voltage. The value of the

Figure 3.12 Phase shift vs applied bias voltage relation


100

80

60

40

20

0

Pha

se S

hift

[deg

.]

10987654321Bias Voltage [V]

Simulation Measured

f = 5.8 GHz

31

beam directions can be calculated by using Eq. (3.1). In Figure 3.12, both

simulation and measured plots are upward that means the phase shift value

(i.e. beam direction) changes with the increase of the bias voltage.

3.3.2 Magic-T

Integrated magic-T structures have been widely used in microwave circuits

and systems. The magic-Ts are four-port devices that offer in-phase and anti-

phase signal division between their two output ports.

Figure 3.13 shows the basic structure of the magic-T used in the proposed

antenna. The magic-T is a combination of a microstrip line T junction and a

slot- to-microstrip line T branch. The signals fed from Port M3 and M4 with

the same phase are combined and the combined signal emerges at Port M1,

as the microstrip line T junction is a parallel branch. On the other hand, as the

slot-to-microstrip line T branch is a series branch, the signals fed from Port

Figure 3.13 Schematic structure of the magic-T.

Port M3

Port M4

Port M1Port M2

Slot Line

Anti-Phase Signal

In-Phase Signal

Microstrip Line

32

M3 and M4 with anti-phase are combined and it emerges at Port M2. Port M1

and M2 are isolated due to the difference of the propagation modes of the

microstrip line and slot line. However, the sum () and difference (Δ) of the

received signals are respectively obtained at Port 1 and Port 2 in Figure 3.6

because the two microstrip antennas are fed from the opposite side of the

patches.

(a) Top View

(b) Bottom View

Figure 3.14 Prototype of the 5.8-GHz beam tracking antenna

(size: 198 × 131 mm) [Reproduced courtesy of The

Electromagnetics Academy].

33

3.4 Prototype E-plane Beam Tracking Antenna

Figure 3.14 shows top view and bottom view photographs of the prototype

5.8-GHz beam tracking antenna. A Teflon fiber substrate (r = 2.15, thickness

= 0.8 mm) is used in the design. The design center frequency is 5.8 GHz and

the size of the proposed antenna is 198 × 131 mm. The patch size is 17.16 ×

17.16 mm. The separation of the antenna elements is 0.8 (=41.4mm). GaAs

tuning varactor (MA46580) is used in the phase shifter. The bias voltage of

each phase shifter is applied via a probe at the center of the patch as shown in

Figure 3.14.

3.5 Result and Discussion

Experiment was done by using a network analyzer (HP8510C) in an anechoic

chamber as shown in Figure 3.15. A standard horn antenna is used as a

transmission antenna. The proposed beam tracking antenna is placed at

receiving side.

Figure 3.15 Experimental set up at anechoic chamber.

34

Figures 3.16 and 3.17 show the return loss plots of Port 1 () and Port2 ()

of the fabricated 5.8-GHz beam tracking antenna. Return losses of different

voltages are shown in these plots. Better than 10-dB return loss is obtained at

5.8 GHz at both Port 1 and Port 2 regardless of the bias voltage.

Figure 3.16 Measured return loss at Port 1[Reproduced courtesy

of The Electromagnetics Academy].

Figure 3.17 Measured return loss at Port 2 [Reproduced courtesy

of The Electromagnetics Academy].

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]

7.06.56.05.55.04.5

Frequency [GHz]

1 V-1 V 1 V-5 V 1 V-9 V 5 V-1 V 9 V-1 V

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]

7.06.56.05.55.04.5

Frequency [GHz]

1 V-1 V 1 V-5 V 1 V-9 V 5 V-1 V 9 V-1 V

35

Figures 3.18 and 3.19 show the measured radiation patterns of and Δ

signals, respectively. The phase shifter is adjusted by changing the bias

voltage of the varactor diodes. In this experiment, bias voltage is applied to

the Right Phase Shifter (RPS) and increased from 1 V to 5 V and 9 V, where

the bias voltage of the Left Phase Shifter (LPS) is fixed to 1 V. As shown in

these figures, the beam of the signal tilts to the right by increasing the bias

Figure 3.18 Measured gain at Port 1 [Reproduced courtesy of The


Figure 3.19 Measured gain at Port 2 [Reproduced courtesy of The


-40

-30

-20

-10

0

Re

lativ

e P

ower

[dB

]

-90 -60 -30 0 30 60 90

Angle [deg.]

1 V-1 V 1 V-5 V 1 V-9 V 5 V-1 V 9 V-1 Vf = 5.8 GHz

-40

-30

-20

-10

0

Re

lativ

e P

ower

[dB

]

-90 -60 -30 0 30 60 90

Angle [deg.]

1 V-1 V 1 V-5 V 1 V-9 V 5 V-1 V 9 V-1 Vf = 5.8 GHz

36

voltage of RPS. Null of the Δ signal also shifts to the right. On the other hand,

when the bias voltage of LPS is increased and that of RPS is fixed to 1 V, the

beam shifts to left as well as null of the Δ signal. In the proposed configuration,

the Δ signal is used to determine the phase shift, i.e. angle of arrival. High

detection accuracy is expected because the null of the Δ signal is sensitive to

the angle of arrival as shown in Figure 3.19.

Table 3.1 shows a comparison between calculated and experimentally

measured results of the relation between beam direction and bias voltage. The

calculated beam direction has been obtained from the simulated result of the

5.8 GHz phase shifter. As shown in Table 3.1, the beam shifts to right 12° and

15°, when the applied bias voltages are 5 V and 9 V, respectively. In this

scenario only RPS bias voltage has changed. On the other hand, when LPS

bias voltage changes to 5 V and 9 V, beam tilts to 12° and 15°, respectively.

The beam shift was 3° when the applied bias voltage was 1 V in both

simulation and experimental result. From the comparison of the data, it is

observed that the measured result is almost similar to the calculated data and

it fulfilled the basic theoretical concept of the proposed antenna.

Table 3.1 Comparison between calculated and measured beam

directions

Bias

Voltage

Beam Direction, (deg.)

Calculated Measured

RPS LPS

1 V 0 3 3

5 V 11.62 12 12

9 V 18.02 15 15

37

Figure 3.20 shows the measured antenna gain with respect to the beam

angle where the bias voltages of the phase shifter is adjusted. The gain

variation over different beam angles is less than 0.7 dB.

Figure 3.21 shows a comparison of the beam shift characteristics of the 5.8-

and 10-GHz antennas. The result is intended by calculating beam direction

Figure 3.20 Measured gain with respect to the beam angle [Reproduced

courtesy of The Electromagnetics Academy].

Figure 3.21 Comparison of the beam shift characteristics of 5.8- and

10-GHz antennas [Reproduced courtesy of The Electromagnetics

Academy].

7.0

6.5

6.0

5.5

5.0

Gai

n [d

Bi]

-20 -15 -10 -5 0 5 10 15 20

Beam Angle [deg.]

f = 5.8 GHz

20

15

10

5

0

Be

am D

irect

ion

[Deg

.]

10987654321

Bias voltage [V]

5.8 GHz 10 GHz

38

from Eq. (3.1), where phase shift value is obtained from the simulation result

of the phase shifters for two different frequencies. These results are consistent

with measured results. The beam shifting operation of the 5.8-GHz antenna

is larger than the 10-GHz beam tracking antenna as the same varactor diodes

are used in the phase shifters for the both frequencies.

3.6 H-plane Beam Tracking Antenna Design

By using the antenna structure of Figure 3.6, the beam tracking in E-plane

can be achieved. To achieve H-plane beam tracking, this section proposes a

new antenna structure.

3.6.1 Operating Principle

Beam tracking operation is same as E-plane beam tracking antenna. The beam

tracking operating principle is discussed in section 3.2. Figure 3.22 shows two

configuration of magic-T integrated antennas. The configuration A and B as

shown in Figures 3.22(a) and 3.22(b) can provide the beam tracking function

in E-plane and H-plane, respectively. In case of E-plane beam tracking, two

antenna elements are fed from the opposite side of the patches as shown in

Figure 3.22(a). Thus signal can be obtained from slot line and signals

from the microstrip line. Therefore and signals are obtained from Port 1

and Port 2, respectively as shown in Figure 3.6. On the other hand, for H-

plane beam tracking antenna is designed such as two antenna elements are

fed from the same side as illustrated in Figure 3.22(b). As antenna elements

are fed from the same side combined in phase signal output from magic-T is

39

obtained from the microstrip line and combined antiphase signal can be

obtained from the slot line. Therefore, signal can be obtained from

microstrip line and signals from the slot line.

(a) Configuration A. Two antenna elements are fed from the

opposite side.

(b) Configuration B. Two antenna elements are fed from same side.

Figure 3.22 Configuration of magic-T integrated antennas.

Port 1

Port 2

Slot LineMicrostrip

Line

Magic-T

Slot Line

Port 1

Port 2

Microstrip Line

Magic-T

40

3.6.2 Antenna Structure

Figure 3.23 shows the structure of the proposed beam tracking antenna for

H-plane. This antenna consists of two phase shifters for phase shifting

operation and magic-T. As the two antenna elements are fed from the same

side of the patches, signal can be obtained from microstrip line and signals

from the slot line of the magic-T. Therefore and signals are obtained from

Port 1 and Port 2, respectively. Thus, changing the phase shift value, an H-

plane beam tracking operation can be achieved.

Figure 3.23 Antenna structure of the H-plane beam tracking antenna


Magic-T

Phase Shifter

Slot Line

Microstrip Antenna

Port 1

Port 2

A A’

A A’

x

x

y

z

y

z

41

(a) Top View

(b) Bottom View

Figure 3.24 Prototype of the 5.8-GHz H-plane beam tracking

antenna (size: 110 110 mm).

42

3.6.3 Prototype Antenna

Figure 3.24 shows the top view and bottom view of the prototype structure

of the H-plane beam tracking antenna. The designed center frequency is 5.8

GHz and the size of the proposed antenna is 110 110 mm. The patch size is

17.12 × 17.12 mm. The separation of the antenna elements is 0.8 (=41.4mm).

GaAs tuning varactor (MA46580) is used in the phase shifter. The bias

voltage of each phase shifter is applied via a probe at the center of each patch.

3.6.4 Result and Discussion

Figure 3.25 shows measured reflection coefficient plots of Port 1 of the

fabricated H-plane beam tracking antenna. In Figure 3.25(a), Left Phase

Shifter (LPS) voltage is fixed to 1 V and Right Phase Shifter (RPS) applied

voltage is varied from 1 V to 9 V. In Figure 3.25(b), LPS voltage is varied

from 1 V to 9 V and RPS voltage is fixed to 1 V. From both figures, it is

observed that the reflection coefficients in all applied voltage conditions are

less than -10 dB at the designed frequency of 5.8 GHz.

Figure 3.26 illustrates the measured return loss plots for Port 2. In case of

Figure 3.26(a), reflection coefficients are observed by varying bias voltage of

RPS. On the other hand, by varying LPS bias voltage return loss of the

proposed antenna is observed and illustrated in Figure 3.26(b). At design

frequency 5.8 GHz, less than -10 dB reflection coefficients are observed.

Figure 3.27 and 3.28 illustrate the measured radiation patterns of the and

43

Δ signals, respectively. The proposed antenna is placed at the receiver side.

The phase shifter is adjusted by changing the bias voltage of the varactor

diodes. In this experiment, the bias voltage is applied to the RPS and

increased from 1 V to 3 V, 5 V, 7 V and 9 V, where the bias voltage of the LPS

is fixed to 1 V. As shown in these figures, the beam of the signal tilts to the

(a) When LPS voltage change.

(b) When RPS voltage change.

Figure 3.25 Reflection coefficient at Port 1 ().

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]

6.56.05.55.04.5Frequency [GHz]

1V-1V 1V-3V 1V-5V 1V-7V 1V-9V

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]


1V-1V 3V-1V 5V-1V 7V-1V 9V-1V

44

right by increasing the bias voltage of RPS. Null of the Δ signal also shifts to

the right as shown in Figure 3.28(a). On the other hand, when the bias voltage

of LPS is increased from 1 V to 9 V and that of RPS is fixed to 1 V, the beam

shifts to left as well as null of the Δ signal as shown in Figure 3.27 (b) and

3.28(b), respectively. From these plots it is confirmed that H-plane beam

tracking operation is achieved successfully.

(a) When LPS voltage changes.

(b) When RPS voltage changes.

Figure 3.26 Reflection coefficient at Port 2 ().

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]


1V-1V 1V-3V 1V-5V 1V-7V 1V-9V

-60

-50

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffici

ent [

dB]


1V-1V 3V-1V 5V-1V 7V-1V 9V-1V

45

(a)Beam shifts to right by increasing RPS voltage.

(b)Beam shifts to left by increasing LPS voltage.

Figure 3.27 Radiation pattern of signal.

-30

-20

-10

0

10

Gai

n [d

Bi]

-90 -60 -30 0 30 60 90

Angle [Deg.]

1V-1V 1V-3V 1V-5V 1V-7V 1V-9V

f = 5.8 GHz

-30

-20

-10

0

10

Gai

n [d

Bi]

-60 0 60

Angle [Deg.]

1V-1V 3V-1V 5V-1V 9V-1V

f = 5.8 GHz

46

(a) Null of the signal shifts to right by increasing RPS voltage.

(b) Null of the signal shifts to left by increasing LPS voltage.

Figure 3.28 Radiation pattern of signal.

-30

-20

-10

0

10

Gai

n [d

Bi]

-90 -60 -30 0 30 60 90

Angle [Deg.]

1V-1V 1V-3V 1V-5V 1V-7V 1V-9V

f = 5.8 GHz

-30

-20

-10

0

10

Gai

n [d

Bi]

-90 -60 -30 0 30 60 90

Angle [Deg.]

1V-1V 3V-1V 5V-1V 7V-1V 9V-1V

f = 5.8 GHz

47

Figure 3.29 Beam angle vs applied voltage plot.

Figure 3.30 Gain vs beam angle plot.

-20

-10

0

10

20

Bea

m A

ngle

[Deg

.]

-10 -8 -6 -4 -2 0 2 4 6 8 10

Applied Voltage [V]

f = 5.8 GHz

10

8

6

4

2

0

Gai

n [d

Bi]

-20 -15 -10 -5 0 5 10 15 20

Beam Angle [Deg.]

f = 5.8 GHz

48

Figure 3.29 explains the relation between beam angle and applied bias

voltage. The data are collected from the measured radiation pattern of the

signal of H-plane beam tracking antenna. The positive value of applied

voltage and beam angle defines when beam shifts to right. Conversely, the

negative value of applied voltage and beam angle states that beam tilts to left.

When 1V is applied to both phase shifters the beam remains at 0. The beam

shifts to right at 3, 6, 9 and 18 when the applied voltage of RPS changes

from 1V to 3V, 5V, 7V and 9V, respectively. Then again, while LPS bias

voltage changes to 3V, 5V, 7V and 9V, beam shifts to left by 6,9,12,15.

Figure 3.30 illustrates the plot of antenna gain with respect to beam angle

where the bias voltages of the phase shifter is adjusted. While beam remain

on 0, the gain is 7.3 dBi. This is the highest gain. The gain decreases when

beam shifts to right and left.

49

3.7 Summary

A new beam tracking concept has been proposed and a 5.8-GHz prototype

beam tracking antenna has been designed, fabricated and measured. The

proposed concept has been successfully confirmed by the theoretical,

simulated and measured results of the proposed antenna. By using a magic-T,

the beam tracking function can be achieved in a simple configuration suitable

for a planar antenna. A different configuration antenna has been also designed

to accomplish the beam tracking in H-plane. H-plane beam tracking antenna

is also fabricated and experimental result is verified. The proposed antenna

can be used for a wide variety of applications in wireless communications and

radar technology.

50

Chapter 4

DOA Estimation Antenna

This chapter proposes a novel planar direction-of-arrival (DOA) estimation

antenna. The estimation capability of phase monopulse DOA estimation

antennas is enhanced by integrating an RF multiplier that detects the phase

relation between the sum and difference of the two received signals. So the

proposed antenna provides a wide range of estimation whereas the

conventional monopulse DOA estimation antennas determine the angles of

half space. A prototype antenna has been fabricated and the proposed concept

was successfully confirmed.

4.1 Introduction

In recent years, due to the improvement of wireless communication

technology, mobile communication has become widespread in general,

further advanced functions are being advanced. In the mobile communication,

the gain greatly changes depending on the arrival angle of the radio wave. In

order to perform higher quality wireless communication, there is an

increasing demand for a compact radio wave direction-of-arrival estimation

device.

There are several types of angle measurement techniques, such as beam

switching method and phased array method. In the beam switching method,

angle measurement is performed by switching fixed beams. In the phased

array system, a plurality of antennas and a transmitter connected thereto are

51

used to change the shape of the beam. These systems become large sized and

not suitable for mounting on a moving body. There is another method named

monopulse for angle measurement method. In the monopulse method, one

signal is received by two antennas, and the angles are measured using the sum

and difference values (Σ and Δ) of the received signals. Since the monopulse

method is an angle measurement method that can easily be downsized

compared with the two principles based on its principle.

There are a number of problems with the arrival angle estimation device in

mobile units. First of all, it has size and weight. Large or heavy items are not

suitable for moving objects. Therefore, miniaturization is a problem. What is

listed next is performance. Because we measure the angle accurately in a

wider range, it becomes a problem to achieve high performance. And many

arrival angle estimation devices are complex and expensive. In order to

realize it at low cost, simplification of the configuration becomes a problem.

In this research, we aim to realize an arrival angle estimation device with a

compact, high performance and simple configuration by using both-sided

circuit technology for a microstrip antenna and constructing an arrival angle

estimation device on the same board.

Many research activities were presented on monopulse antennas to enrich

the research field in this topic. A monopulse microstrip antenna array in a

single layer was proposed to achieve two dimensional monopulse

performance [45]. Four 3-dB hybrid couplers and several 90 delay lines are

used as a comparator in this design. A multilayer 44 microstrip patch antenna

was introduced to broaden the impedance bandwidth of the antenna in C-band

52

[46]. Another monopulse antenna was designed by integrating an

electromagnetically coupled (EMCP) microstrip antenna array structure and

used a rat-race hybrid instead of a branch line coupler as a comparator so that

it can be used as a transmitter in addition to a receiver [47]. A dual probe fed

single aperture monopulse antenna was presented to eliminate complex

phasing networks and aperture arrangement of the antenna. This structure

consists of a slot embedded patch, two probe inputs and a 180 directional

coupler [48]. The DOA estimation concept based on monopulse mechanism

has been already discussed [19]. The classical monopulse concepts are

extended to a general complex monopulse concept which can utilize

information from side lobes by using phase shift of the signals.

Previous researchers in our laboratory also proposed planar DOA

estimation antennas based on the phase monopulse mechanism [36, 37].

These antennas are based on the conventional phase monopulse mechanism

where only the amplitude of the sum () and difference (Δ) of the two

received signals are evaluated. Therefore, these antennas detect the arrival

angle in only the half plane of the space. In order to achieve wide range of

estimation angles, an extended DOA estimation antenna is proposed in this

chapter which determines the phase relation between and Δ signals as well

as there amplitudes. This antenna is integrated with RF multiplier and DOA

operation is successfully achieved.

In this chapter, a novel 10-GHz DOA estimation antenna integrating an RF

multiplier is proposed to enhance the estimation capability of arrival angles.

This antenna consists of microstrip lines and slot lines on both sides of the

53

substrate and microwave circuits such as a magic-T and RF multiplier are

integrated with the antenna elements. The proposed antenna has a wide range

of the DOA estimation capability by effectively using the phase relation of

the and Δ signals as well as their amplitudes. A prototype antenna was

designed, fabricated and experimentally evaluated.

The configuration and basic operation principle of the proposed DOA

estimation antenna are discussed in section 4.2. In section 4.3, the structure

and design of the antenna which integrates microstrip antennas, magic-Ts, an

RF multiplier and power dividers are described. Section 4.4 demonstrates a

prototype DOA estimation antenna designed at the 10-GHz frequency band.

The measured results of the radiation pattern of the and Δ signals and DOA

estimation performance are highlighted in this section. Finally, section 4.5

concludes this chapter.

4.2 Operating principle

Figure 4.1 illustrates a basic block diagram of the proposed extended

monopulse DOA estimation antenna composed of two antennas, a magic-T,

an RF multiplier and two detectors. As magic-Ts provide in-phase or anti-

phase power division according to the input and output ports, the sum () and

difference (Δ) of the signals received by the two antennas are separately

obtained. The RF multiplier is used to detect the phase relation between the

and Δ signals and it provides sign of the Δ signal. The direction of the

received arrival wave, i.e. plus or minus of the arrival angle , depends on

54

this sign of the Δ signal.

Figure 4.2 describes the basic operating principle of the proposed extended

monopulse DOA estimation antenna with vector diagrams. A radio wave

arriving from an angle is received by two antenna elements #1 and #2 with

phase difference . In the vector diagrams, the blue and purple arrows show

the signals received by the antenna elements #1 and #2, respectively. The red

and green arrows show the and Δ signals, respectively. In the scenario of

Figure 4.2(a), the radio wave comes from the left side of the antennas. In this

case, the arrival angle is defined as negative. On the other hand, is defined

Figure 4.1 Block Diagram of the proposed DOA estimation antenna

concept. Sum () and diference () of the two received signals are

generated by a magic-T. Sign of signal can be obtained from an RF

multiplier.

Magic- T

RF Multiplier

Arrival Wave

#1 #2

Sign()

55

as positive in the scenario of Figure 4.2(b).

When two signals received by the two antenna elements #1 and #2 have

different phase with the same amplitude, sum () and difference () of the

received signals can be expressed by the following equations by assuming the

two antenna elements to be isotropic:

2cos2

eeΣ 22

A

AAjj

(4.1)

(a) When < 0, Δ is advanced from

(b) When > 0, Δ lags

Figure 4.2 Basic operating principle of the proposed DOA estimation

antenna [Reproduced courtesy of The Electromagnetics Academy].

Arrival Wave

#1 #2

dsin

Im

Re

#1

#2

2

Im

Re

#2

#1

Arrival Wave

#1 #2

dsin

2

56

2sin2

eeΔ 22

jA

AAjj

(4.2)

where A and ( 21 ) are the amplitude and phase difference between

two received signals, respectively.

From the vector diagram shown in Figure 4.2 and Eqs. (4.1) and (4.2),

the phase monopulse DOA estimation principle can be explained. The phase

difference is expressed as following equation:

Σ

Δ

2tan

Σ

Δtan2 1 (4.3)

The phase monopulse mechanism can relate the arrival angle () and phase

difference () of the two received signals by following expression:

dθ

π2

λsin 1 (4.4)

where d is the antenna separation and is the wavelength. Thus the

monopulse mechanism can be expressed by the following Eq. (4.5), where

the amplitude of the sum (||) and difference (||) of the two received signals

are used to determine the arrival angle [36-38].

Σ

Δtan

π

λsin 11

dθ (4.5)

Conventional monopulse DOA estimation antennas determine the arrival

angle by only the amplitude of and Δ signals. The phase relation between

and Δ signals is not considered as well as the sign of Δ signal. As a result,

57

conventional monopulse DOA estimation antennas cannot determine the sign

of arrival angles .

The vector diagrams illustrated in Figure 4.2 indicate two different

scenarios. From the vector diagrams and Eqs. (4.1) and (4.2), it is clarified

that the phase difference between and Δ signals is always 90. As shown in

the vector diagrams, the phase relation between two received signals can be

determined by calculating the phase relation between and Δ signals. Figure

4.2(a) shows #1 leads #2 and Δ is advanced from . The arrival angle is

negative in this scenario. On the other hand, is positive and Δ lags in the

scenario of Figure 4.2(b). Thus the antenna can determine the arrival angle on

the both sides by evaluating the phase relation between and signals.

An RF multiplier is employed in this structure to determine whether the

phase of Δ signal is advanced or delayed from signal. When the and

signals are input from two RF inputs of the RF multiplier, the output DC

voltage VMul can be expressed by following equation:

cosMulV (4.6)

where Σ and Δ are the phase of the two input signals and ,

respectively [49]. As shown in Figure 4.2 and Eqs. (4.1) and (4.2), and

signals always have 90 phase difference. Therefore the feed circuit between

the magic-T and RF multiplier has to be designed to give additional 90 phase

difference between and signals so as to ΔΣ has a value of 0 or 180.

When the signal inputs are in phase, i.e., 0ΔΣ , a positive voltage is

obtained as VMul. On the other hand, if the input signals are in opposite phase,

i.e., 180Δ , a negative voltage is obtained. This makes it possible to

58

discriminate between in-phase and anti-phase based on the positive and

negative of the DC voltage output. Depending on the sign of the output

voltage, the direction of the received arrival wave can be perceived.

4.3 Antenna Structure

Figure 4.3 shows the structure of the proposed antenna. The proposed antenna

consists of four microstrip antennas, two magic-Ts, two power dividers and

an RF multiplier. RF signals received by the antennas are combined and sum

() and difference () of the received signals are obtained from magic-Ts.

The amplitude of the and Δ signals can be obtained from Port 1 and Port 2,

Figure 4.3 Structure of the proposed planar DOA estimation antenna. Two

magic-Ts and an RF multiplier employing the both-sided MIC technology

are integrated with four antenna elements [Reproduced courtesy of The


RF Multiplier

Microstrip LineMicrostrip Antenna

Magic-T

A A’

A’A

Slot Line Port 2Port 1

xz

φ

x

z

y

y

Power Divider 1

Power Divider 2

59

respectively. Two power dividers are integrated in the proposed antenna

structure to divide power to the RF multiplier and output ports of the and Δ

signals. Sign of the signal is obtained at the inner conductor of the slot ring

of the RF multiplier. Both side MIC technology has been perfectly applied in

this proposed antenna structure [20]. However, it is easy to integrate them in

the proposed structure as presented in [37].

4.3.1 RF Multiplier

This section will describe the configuration, operating principle, design and

characteristics of the RF multiplier used for wide angle estimation of DOA

estimation antenna structure. Here, the principle of an RF multiplier will be

described. Figure 4.4 illustrates the structure of the RF multiplier. It consists

of two microstrip lines, a slot line, a ring slot and two diodes. Output voltage

of this multiplier, obtained from DC port, depends on the phase difference of

the input signals from RF 1 and RF 2 as shown in figure.

Figure 4.4 Structure of the RF multiplier. Sum () and difference () of

the two received signals are fed to Port RF1 and RF2 respectively. By

observing the output voltage, sign of signal can be perceived


D1

D2

Slot Ring Slot Line

Output Voltage

Microstrip Line

Diode

Port RF2

Port RF1

DC Port

60

Equations can be derived from diode’s nonlinear equation. A typical

nonlinear element is a PN junction diode characteristic is approximated by

the following equation.

1exp0

nkT

qVII (4.7)

In the above equation, I0 is reverse saturation current, q is the unit charge,

V is the bias voltage, n is the ideal coefficient, usually between 1 and 2 and k

is the Boltzmann constant, and T is the absolute temperature.

Consider current flowing in one diode as shown in Figure 4.5. From

equation (1) diode current I can be described as follows.

1exp0

nkT

qVII

The above expression can be expressed to Maclaurin series

V)nkT

qexp(I)

nkT

q(

V)nkT

qexp()I

nkT

q(

2

0

1

00

02

2

2

0

2

dV

Id)V(''I

dV

dI)V('I

V!

)(''IV

!

)('I)(I)V(I

(4.8)

Figure 4.5 Two diode balanced multiplier.

I1 I2ID

V1

V2

V1

V2

D1 D2

61

Here, since we use the nonlinear part of the diode, consider the term of the

square and let that term be ID1,

20

221 )(

2

1

2

0VI

nkT

qV

!

)(''IID (4.9)

Voltage across diode can be shown by following equation:

)sin()sin( 22111 tVtVVD (4.10)

By squaring the expression (4.10) and transforming it using the equations

(4.11) and (4.12), it becomes the equation (4.13):

sin . sin cos cos (4.11)

sin 1 cos 2 (4.12)

12

1 cos 2 1 cos 2

cos 2 cos (4.13)

Substituting equation (4.13) to equation (4.9) the following equation can

be achieved

. 1 cos 2 1 cos 2

2 . cos 2 cos (4.14)

Here,

4

As shown in Figure 4.5, the currents flowing through D1 and D2 are set,

and the voltage is set such that the sum of the two signals is applied to D1 and

the difference between the two signals is applied to D2. Therefore, the AC

voltage V applied to the diode of the formula (4.10) is set as follows. .

sin sin

sin sin

62

Current ID1 and ID2 can be expressed by following equation

. 1 cos 2 1 cos 2

2 . cos 2 cos

. 1 cos 2 1 cos 2

2 . cos 2 cos

So the diode current ID can be expressed by the following equation:

4 . cos 2 cos

By ignoring the RF component part, the above equation can be expressed

as following:

)cos( 2121 VVID

Therefore, the output voltage from the central conductor of the RF

multiplier can be described as following

)cos( 2121 VVIV DD

When and signals are input from RF 1 and RF 2, output of RF

multiplier VMul can be achieved by following equation.

)cos(ΔΣMul .V

Here and are the amplitude of the and signal and φ and φ are

their phase, respectively.

In this equation when 0 , VMul is positive and when 180 ,

63

VMul is negative. Thus the antenna is designed such as the phase difference

between this signals remain either 180 or 0 to get either positive or negative

value of the RF multiplier output.

By replacing the amplitude of sum and difference from Eqs. (4.1) and (4.2)

and phase difference () from Eq. (4.4), RF multiplier output and the arrival

angle () can be relate by the following expression:

∝ |Σ|. |Δ| cos

∝ 4 cos sin sin sin cos (4.15)

Figure 4.6 illustrates the relationship between output VMul and arrival angle

, derived from Eq. (4.15). Antenna is assumed as isotropic. Negative bias

voltage achieved when > 0 and positive output achieved when < 0.

Antenna element separation is assumed as 0.8 of the wavelength.

Figure 4.6 Theoretical relation between the RF multiplier output

voltage and arrival angle [Reproduced courtesy of The


-0.6

-0.4

-0.2

0.0

0.2

0.4

0.6

RF

Mul

tiplie

r O

utpu

t [a.

u.]

-40 -30 -20 -10 0 10 20 30 40Arrival Angle [deg.]

d = 0.8

64

4.3.2 Power Divider:

The proposed extended monopulse DOA estimation antenna integrates two

power dividers to divide the power from the magic-Ts to the RF multiplier

and the output port of the and signals. Power dividers are designed to

provide enough power to the output ports and the remaining power to the RF

(a) Power divider 1. This power divider is used to divide the power of

signal.

(b) Power divider 2. This power divider is used to divide the power of

signal.

Figure 4.7 Two input two output power dividers used to divide the

powers to RF ports and RF multiplier [Reproduced courtesy of The


P1

P3 P4

P2

Microstrip Line

154

50

50

31.25

P5

P6

P7 P8

Microstrip LineSlot Line

100

100

125 50

65

multiplier because the RF multiplier only determines the phase relation of the

two input signals.

Figure 4.7 illustrates two power dividers, power divider 1 and power

divider 2. Power divider 1 in Figure 4.7(a) is a two way power divider which

consists of microstrip lines. Port impedances are designed as 50 , 50 , 154

and 31.25 for P1, P2, P3 and P4, respectively. On the other hand, power

divider 2 as shown in Figure 4.7(b) is formed by microstrip lines and slot lines.

100- port impedance is designed for Port P5 and P6. Port impedances for

P7 and P8 are designed as 50 and 125 , respectively. Here, the input

signals of Port P1 and P2 are the signals of the magic-T networks. In case

of the power divider 2, two signals from magic-Ts are input to Port P5 and

P6 and output from Port P7 and P8.

Figure 4.8 shows the simulated performance of the power dividers. As

illustrated in Figure 4.8(a), reflection coefficient is less than -10 dB for all the

input ports P1, P2, P5 and P6. According to Figure 4.8(b), at power divider 1,

around -10 dB of the input power is transferred from P1 and P2 to P3 and

around -4.3 dB goes to P4 from P1 and P2. Almost similar characteristics are

observed for the power divider 2 where the amount of power transferred from

P5 and P6 to P7 and P8 are around -5.5 dB and -9.6 dB, respectively.

66

(a) Power divider 1. This power divider is used to divide the power of

signal.

(b) Power divider 2. This power divider is used to divide the power of

signal.

Figure 4.8 Two input two output power dividers used to divide the

powers to RF ports and RF multiplier [Reproduced courtesy of The


-20

-15

-10

-5

0

Ref

lect

ion

Coe

ffic

ient

[dB

]

11.010.510.09.59.0

Frequency [GHz]

S11, S22

S55, S66

-20

-15

-10

-5

0

Tra

nsm

issi

on

Co

effi

cie

nt

[dB

]

11.010.510.09.59.0Frequency [GHz]

S31, S32

S41, S42

S75, S76

S85, S86

67

4.4 Measured Result


Figure 4.9 shows the top and bottom view of the prototype 10-GHz extended

monopulse DOA estimation antenna. In this design, a Teflon glass fiber (r =

2.15, thickness = 0.8 mm) is used as a substrate material. The size of the

proposed antenna is 113113 mm and the patch size is 9.659.65 mm. The

(a) Top view

(b) Bottom view. The RF multiplier output is acquired from this wire.

Figure 4.9 Prototype 10-GHz DOA estimation antenna (Size: 113113

mm, f =10 GHz) [Reproduced courtesy of The Electromagnetics

Academy].

68

antenna element separation is 0.8 (=24mm) as the designed center frequency

is 10 GHz. Two Schottky diodes (Metelics, MSS30,154-B10B) are used for

the RF multiplier. The RF multiplier output can be obtained from the wire

connected to the inner conductor of the slot ring as shown in Figure 4.9(b).

4.4.2 Result and Discussion

Figure 4.10 shows the measured reflection coefficient plots of Port 1 and

Port 2 of the designed antenna. Better than 10-dB return loss is observed at

9.85 GHz for both ports. The impedance bandwidth of Δ signal is 1.52% and

the impedance bandwidth of 6.94% is obtained for signal. Though the

antenna has been designed at 10 GHz, the measured result shows that the

minimum reflection coefficient is observed at 9.85 GHz.

Figure 4.11 illustrates the simulated and measured radiation patterns of

and Δ signals. Simulation is done by using Keysight Technologies’ ADS

simulation software and diodes are not attached in the simulation process. The

simulated gain of this antenna is 12.3 dBi. In the experiment, 9.37 dBi

maximum gain is achieved for this antenna.

Figure 4.12 presents the theoretical and measured data of the relation

between the arrival angle and |Δ|/||. Theoretical data are calculated from Eq.

(4.15) and measured data are obtained from calculating the measured gain.

From this Figure, it is obtained that the range of arrival angle is -38 to 38

for theory and -42 to 39 for measured data.

69

Figure 4.10 Measured reflection coefficient plots of Port 1 and Port 2

of the designed antenna [Reproduced courtesy of The


Figure 4.11 Simulated and measured radiation patterns of and Δ

signals [Reproduced courtesy of The Electromagnetics Academy].

-40

-30

-20

-10

0

Ref

lect

ion

Coe

ffic

ient

[dB

]

11.010.510.09.59.0Frequency [GHz]

S11

S22

-30

-15

0

15

Gai

n [d

Bi]

-180 -120 -60 0 60 120 180Angle [deg.]

(Sim.) (Sim.) (Exp.) (Exp.)

f = 9.85 GHz

70

Figure 4.12 Relation between / and arrival angle . Arrival angle

can be determined from this plot and sign of the signal [Reproduced

courtesy of The Electromagnetics Academy].

-30

-20

-10

0

10

20

30

/[

dB]

-90 -60 -30 0 30 60 90

Angle [deg.]

Measured Theory

Figure 4.13 Experimental setup in anechoic chamber to measure output

of RF multiplier.

71

Figure 4.13 illustrates the experimental setup to measure the output of the

RF multiplier in an anechoic chamber. A horn antenna is used as a

transmitting antenna. The proposed DOA estimation antenna integrated with

RF multiplier is placed 71 cm distance from the transmitting antenna.

Frequency of the transmitting antenna is varied from 9.75 GHz to 10.2 GHz

and output of the RF multiplier is measured by digital multimeter (Agilent

34401A).

Figure 4.14 shows the output voltage of the RF multiplier in eight different

frequencies. When the arrival angle is greater than zero, the output voltage is

negative. When the arrival angle is less than zero, the output voltage is

positive. Depending on the sign of this output voltage, sign of the signal

can be determined. In case of 10.2 GHz, the polarity has become opposite. It

may be caused due to the phase difference obtained at the input signals of the

RF multiplier. The phase may change because of the different wavelengths

for different frequency signals.

Figure 4.15 shows the RF multiplier output voltage at four different power

densities 0.23, 0.14, 0.09 and 0.06 W/m2. The RF multiplier works even at

the power densities as low as 0.06 W/m2.

72

Figure 4.14 Measured RF multiplier output voltage vs. arrival angle

(Frequency dependence) [Reproduced courtesy of The


Figure 4.15 Measured RF multiplier output voltage vs. arrival angle

(Power density dependence) [Reproduced courtesy of The


-90

-60

-30

0

30

60

90

RF

Mul

tiplie

r O

utp

ut [

mV

]

-45 -30 -15 0 15 30 45Arrival Angle [deg.]

9.75 GHz 9.8 GHz 9.85 GHz 9.9 GHz 9.95 GHz 10 GHz 10.1 GHz 10.2 GHz

PD = 0.23 W/m2

-90

-60

-30

0

30

60

90

RF

Mul

tiplie

r O

utp

ut [

mV

]

-45 -30 -15 0 15 30 45Arrival Angle [deg.]

0.23 W/m2

0.14 W/m2

0.09 W/m2

0.06 W/m2

f = 9.85 GHz

73

4.5 Summary

This chapter has presented an RF multiplier integrated planar antenna for

extended monopulse DOA estimation. The proposed antenna enhances the

estimation range of the arrival angle by evaluating the phase relation between

the sum and difference signals, whereas the half angle of the space can be

estimated by conventional DOA estimation antennas. The performance of the

RF multiplier and radiation characteristics of the antenna have been

experimentally verified. Simple structure makes the proposed antenna

attractive for a wide range of applications in DOA estimation.

74

Chapter 5

Dual Axis DOA Estimation Antenna

A novel multilayer structure of DOA estimation antenna is proposed in this

chapter. This antenna provides dual plane angle of arrival estimation whether

conventional DOA estimation antenna determine arrival angle in single plane.

The proposed antenna has a multilayer structure and consists of four ring slot

antennas as antenna elements and four magic-Ts. The proposed antenna

employs monopulse mechanism and provides both xz-plane and yz-plane

DOA estimation.

5.1 Introduction

In wireless communication system, it is extremely crucial to estimate the

direction of incoming signals in order to achieve better reception. Direction

of arrival (DOA) estimation for has been investigated extensively in the last

six decades and has been applied in various fields including wireless

communication, radar, sonar and audio processing [50-53]. The role of DOA

estimation in wireless communication is essential since it helps to estimate

the direction of the incoming signal. The estimated signal direction will be

used to point the array beam towards the estimated direction.

DOA estimation of the radio wave has been an interesting area of research

as it offers several interesting benefits in terms of improved Quality-of-

Service, such as, better coverage, more reliable communication, and higher

data rates [54]. Furthermore, the DOA information can also be used for

75

positioning or localization in a wireless cellular network.

DOA antenna using magic-T was already proposed [36]. This DOA

estimation antenna can determine the angle of arrival in single plane by

employing monopulse mechanism. In this chapter, a new dual-axis DOA

estimation antenna is described which can calculate the angle of arrival

signals in two planes. It has a multilayer structure and four ring slot antennas

are used as antenna elements.

Firstly, section 5.2 describe the basic operating principal of the proposed

dual-axis DOA estimation antenna. Details of the antenna structure are

described in section 5.3. Simulation and experimental results are described in

section 5.4. Finally section 5 conclude the chapter.

76

Figure 5.1 Block operating principle of the DOA estimation Antenna

concept.

Figure 5.2 Concept of the dual-axis DOA estimation antenna.

dy

Arrival wave

∑yz

∆yz

Magic-T1

dx ∑xz

∆xz

xy

z Magic-T2

77

5.2 Basic Operation Principle

Figure 5.1 illustrates the basic block diagram of the proposed antenna. Three

antenna elements are arranged along the x- and y-axis with the antenna

separation of dx and dy. The antenna elements along the x- and y-axis connect

to Magic-T1 and -T2, respectively. As the magic-T provides in-phase or anti-

phase power combination according to the input and output ports, the sum ()

and difference () of the signals received by the two antenna elements are

separately obtained [31]. Therefore, the sum and difference signals in the x-

axis, i.e. xz and xz are obtained from Magic-T1. Similarly, yz and yz are

obtained from Magic-T2. The arrival angle of the received signals can be

determined by the amplitude of the sum () and difference () of the two

received signals using the monopulse mechanism.

Figure 5.2 illustrated the coordinate system where xz and yz are defined

as arrival angles of xz-plane and yz-plane, respectively. In case of two

dimensional DOA estimation, the arrival angle in xz-plane, xz and yz-plane,

yz can be expressed by following equations:

xz

11

Σ

Δtansin xz

x

xz d (5.1)

yz

yz11

Σ

Δtansin

y

yz d (5.2)

where dx and dy are antenna separation in x-axis and y-axis, respectively. is

the wavelength. xz and xz are and signals in xz-plane and yz and yz are

and signals in yz-plane, respectively. Therefore the proposed array

antenna employs monopulse mechanism and provides DOA estimation in

78

both xz-plane and yz-plane.

5.3 Antenna Structure

Figure 5.3 shows the schematic layout of the multilayer dual-axis DOA

estimation antenna. The antenna comprises four ring-slot antenna elements

and four magic-Ts. The complete structure of the antenna is designed in three

metal layers. The ring-slot antennas are formed in Layer 2 and the feed

networks are distributed on two different layers (i.e., Layer 1 and Layer 3).

Two separate feed networks are designed in two separate planes so that the

arrival angle estimation operation can be obtained in two orthogonal planes

(i.e. xz-plane and yz-plane). One of the feed networks with Port 1 and 3, which

respectively provide and signals in the xz-plane is arranged on Layer 1.

Figure 5.3 Schematic structure of the antenna (5.8 GHz).

79

Layer 3 consists of another feed network with Port 2 and 4, which provide

and signals in the yz-plane, respectively. Two slot lines arranged

orthogonally are also formed with four slot rings in Layer 2. In this structure,

Layer 1 detects only horizontal polarization (x-polarization) while Layer 3

detects only vertical polarization (y-polarization).Therefore, the arriving

waves have to have both x- and y-polarization to properly estimate the arrival

angle in the dual-axis. In case of circular polarizations, this antenna works

perfectly.

5.4 Result and Discussion

5.4.1 Simulated Result

The proposed antenna is designed using ADS simulation software. The

simulation results are discussed in this section. Figure 5.4 demonstrates the

simulation result of the return loss of and signals for both layers for port

1, 2, 3 and 4. Better than 10-dB return loss is obtained at design frequency

5.8 GHz.

Figure 5.5 illustrated the radiation pattern plot of and signals in xz-plane

and yz-plane. 9.81 dBi and 9.2 dBi simulated gain are observed for yz and

xz , respectively. Thus, the proposed antenna can determine the arrival angle

in both xz-plane and yz-plane by calculating xz and yz from Eqs. (5.1) and

(5.2).

80

Figure 5.4 Simulated reflection coefficient plot of Port 1, 2, 3 and 4.

Figure 5.5 Simulated radiation pattern plot of and signal in xz-

plane and yz-plane.

-40

-30

-20

-10

0R

efle

ctio

n C

oeffi

cien

t [d

B]

6.56.46.36.26.16.05.95.85.75.65.5Frequency [GHz]

S11

S22

S33

S44

-20.0

-15.0

-10.0

-5.0

0.0

5.0

10.0

Gai

n [d

Bi]

-180 -120 -60 0 60 120 180

Angle [Deg.]

yz

yz

xz

xz

81

(a) Layer 1

(b) Layer 2

(c) Layer 3

Figure 5.6 Prototype of the 5.8-GHz dual axis DOA estimation

antenna. (size: 88mm 88mm)

82


Figure 5.6 shows the photograph of the fabricated 5.8-GHz dual-axis DOA

estimation antenna in three different layers. and signals of the xz-plane

from Port 1 and Port 3 are obtained at Layer 1, and Port 2 and Port 4 at Layer

3 provide and signals of the yz-plane, respectively. In this design, a

Teflon fiber (r = 2.15, thickness = 0.8 mm) is used as a substrate material.

The size of the prototype antenna is 8888 mm. Each ring-slot antenna has

an outer radius 7 mm and inner radius 6 mm resulting a slot width of 1 mm.

5.4.3 Measured Result

The experiment of the proposed antenna have been performed in an anechoic

chamber as shown in Figure 5.7. The proposed antenna is used as receiving

antenna and linearly polarized horn antenna is used as a transmitting antenna

during experimental procedure. During radiation pattern measurement, the

proposed antenna is placed on a rotating table that can rotate 360.

Figure 5.8 shows the measured reflection coefficient plots of Port 1, 2, 3

and 4 of the designed antenna. Better than 10-dB return loss is observed at

5.64 GHz for all ports. Though the antenna has been designed at 5.8 GHz, the

measured result shows that the minimum reflection coefficient is observed at

5.64 GHz for all ports.

Figure 5.9 illustrates the measured radiation patterns of and Δ signals for

xz-plane and yz-plane. The measured gain of this antenna is 6.84 dBi for yz

and 5.32 dBi for xz. It is possible to create very small air gap between two

83

separate layers of the fabricated antenna which might create some

imperfection. So there is a 3 dB gain difference between simulated and

measured gain is obtained. The concept of dual axis DOA estimation can be

confirmed by the experimental result.

Figure 5.7 Experimental orientation in anechoic chamber.

Figure 5.8 Measured reflection coefficient plot of Port 1, 2, 3 and 4.

-40

-30

-20

-10

0

Re

flect

ion

Coe

ffici

ent [

dB]


S11

S22

S33

S44

84

5.5 Conclusion

A multilayer structure of a DOA estimation antenna is discussed in this

chapter. The proposed antenna can determine the arrival angle of the received

signals in two planes. The antenna has been fabricated and the performance

has been measured to confirm DOA estimation operation. It is found that the

antenna can determine the arrival angle at xz- and yz-planes according to the

theory.

Figure 5.9 Measured radiation pattern plot of and signal in xz-

plane and yz-plane.

-40

-30

-20

-10

0

10G

ain

[dB

i]

-180 -120 -60 0 60 120 180

Angle [deg.]

xz

yz

xz

yzf = 5.64 GHz

85

Chapter 6

Conclusion

In this thesis, a beam tracking antenna and DOA estimation antenna concept

are proposed. RF signal processing technology is integrated with planar array

antennas which brings the new model of antenna devices in wireless

communication technology. The proposed structure makes the antenna

structure very simple and compact. Proposed concept of beam tracking and

DOA estimation are verified by experimentally.

Phase shifters are integrated with antenna array elements to adjust the beam

direction in beam tracking antenna. By adjusting the phase shifters, the

antenna beam can shift to the direction of arrival wave. A magic-T is also

integrated in this structure which reduce the complexity of design structure.

The both sided MIC technology is effectively employed to realize the array

antenna. This type of antenna can be used for the applications of radar

tracking, air traffic control system, vehicle tracking, blind spotting, cruise

control etc.

RF multiplier is integrated with the planar antenna array structure for DOA

estimation. Integrating RF multiplier makes the DOA estimation antenna to

improve the estimation capability. By observing the RF multiplier output it is

possible to detect the arrival angle of the received signal in wide space. This

gives the proposed antenna a unique feature than the other conventional DOA

estimation antenna. The proposed antenna performance is successfully

86

confirmed experimentally. DOA estimation antenna is essential and key

component to perform higher quality wireless communication.

To achieve DOA estimation in dual axis, a multilayer structure planar

antenna is also proposed in the thesis. Annular slot antenna arrays are

integrated with microwave integrated circuit magic-T and perform DOA

estimation. Three layers structure of the antenna make a single antenna to

perform DOA operation in two planes. This structure also employs both side

MIC technology. Thus DOA estimation performance can be improved by

proposed antennas.

87

References

[1] C.A. Balanis, “Antenna Theory: Analysis and Design,” 2nd Ed., New

York: John Wiley & Sons, Inc., 1996.

[2] J.D. Kraus, “Antennas,” 2nd Ed., New York: McGraw-Hill, 1998.

[3] J. L. Volakis, “Antenna Engineering Handbook,” 4th Ed., New York:

McGraw-Hill, 2007.

[4] IEEE standard definitions of terms for antennas, IEEE Trans. On

Antennas and Propag., vol. 17, no. 3, pp. 262-269, 1969.

[5] H. Hertz, “Electric Waves,” Mac Millan Co., 1893.

[6] D. G. Fang, “Antenna theory and microstrip antennas,” CRC Press, 2010.

[7] G. A. Deschamps, “Microstrip microwave antennas,” 3rd USAF Symp.

on Antennas, 1953.

[8] R.L. Haupt and M. Lanagan, “Reconfigurable antennas,” IEEE Antennas

and Propag. Mag., Vol. 55, No. 1, 49-61, 2013.

[9] R. E. Munson, “Microstrip phased array antennas,” Proc. Of 22 Symp. On

USAF Antenna Research and Development Program, Oct 1972.

[10] R. E. Munson, “Conformal microstrip antennas and microstrip phased

arrays,” IEEE Trans. Antennas Propagat., vol. AP-22, no. 1, pp. 177-

180, Jan. 1974.

[11] J. Q. Howell, “Microstrip antennas,” IEEE Trans. Antennas Propag., vol.

AP-22, pp. 90-93, Jan. 1974.

[12] A. G. Derneryd, “Linearly polarized microstrip antennas,” IEEE Trans.

Antennas Propagat., vol. AP-24, no. 6, pp. 846-850, Nov. 1976.

[13] Y. T. Lo, D. Solomon, and W. F. Richards, “Theory and experiment on

microstrip antennas,” IEEE Trans. Antennas Propagat., vol. AP-27, no.

3, pp. 137-145, Mar. 1979.

[14] M. D. Deshpande and M. C. Bailey, “Input impedance of microstrip

antennas,” IEEE Trans. Antennas Propag., vol. AP-30, no. 4, pp. 645-

650, Jul. 1982.

88

[15] H. Wang and D. G. Fang, “A compact single layer monopulse microstrip

antenna array,” IEEE Trans. Antennas and Propag. vol. 54, no. 2, 503-

509, 2006.

[16] W. U. Zhong, G. M. Wang, and C. X. Zhang, “A broadband planar

monopulse antenna array of C band,” IEEE Antenna and Wireless

Propag. Letters, Vol. 8, 1325-1328, 2009.

[17] K. Hemant and G. Kumar, “Microstrip antenna array with ratrace

comparator at X-band for monopulse tracking radar,” 2016 IEEE Int’l

Symp. On Antennas and Propag. (APS-URSI), 513-514, Fajardo,

Puerto Rico, 2016.

[18] Y. Fei, Y. Xie, and L. Zhang, “Single patch antenna with monopulse

patterns,” IEEE Microwave and Wireless Comp. Letters, Vol. 26, 762-

764, 2016.

[19] W. Kederer and J. Detlefsen, “Directon of arrival (DOA) determination

based on monopulse concepts,” Proc. 2000 Asia-Pacific Microwave

Conf. (APMC2000), 120-123, Sydney, Australia, 2000.

[20] M. Aikawa and H. Ogawa, “Double-sided MICs and their applications,"

IEEE Trans. Microwave Theory & Tech., Vol. 37, No. 2. Pp. 406-413,

1989.

[21] C. L. Mak, K. M. Luk, and K. F. Lee, “Microstrip line-fed l-strip patch

antenna,” IEEE Proceedings Microwaves, Antennas and Propagation,

vol. 146, no. 4, pp. 282-284, 1999.

[22] K. S. Fong, H. F. Pues, and M. J. Withers, “Wideband multilayer

coaxial-fed microstrip antenna element,” Electronic Letters, vol. 21, no.

11, pp. 497-499, 1985.

[23] I. Park, B. J. Cho, and Mittra R., “An aperture-coupled small microstrip

antenna with enhanced bandwidth,” IEEE International symposium on

Antennas and Propagation, pp. 1212-1215, 1999.

[24] D. Sun and L. You, “A broadband impedance matching method for

proximity-coupled microstrip antenna,” IEEE Trans. on Antennas and

propagation, vol. 58, no. 4, pp. 1392-1397, 2010.

89

[25] C. H. Ho, L. Fan, and K. Chang, “New uniplanar coplanar waveguide

hybrid-ring couplers and magic-ts,” IEEE Transactions on Microwave

Theory and Techniques, Vol. 42, No. 12, 2440–2448, December 1994.

[26] M. Aikawa and H. Ogawa, “A new MIC magic-T using coupled slot

lines,” IEEE Transactions on Microwave Theory and Progress In

Electromagnetics Research, Vol. 129, 2012 107 Techniques, Vol. 28, No.

6, 523–528, June 1980.

[27] C. H. Cheng and Q. Gao, “A novel broadband magic-T using microstrip-

slotline transitions,” Microwave and Optical Technology Letters, Vol.

46, No. 6, 585–588, September 2005.

[28] Y. W. Yu, G. M. Wang, and Y. Ding, “A novel wideband planar magic-

T,” International Conference on Microwave and Millimeter Wave

Technology, ICMMT, Vol. 1, 315–317, 2008.

[29] W. Feng, Q. Xue, and W. Che, “Compact planar magic-T based on the

double-sided parallel-strip line and the slotline coupling,” IEEE

Transactions on Microwave Theory and Techniques, Vol. 58, No. 11,

2915–2923, November 2010.

[30] F. F. He, K. Wu, W. Hong, H. J. Tang, H. B. Zhu, and J. X. Chen, “A

planar magic-T using substrate integrated circuits concept,” IEEE

Microwave and Wireless Components Letters, Vol. 18, No. 6, 386–388,

June 2008.

[31] M. Aikawa and E. Nishiyama, “Compact MIC magic-T and the

integration with planar array antenna,” IEICE Trans. Electron., Vol.

E95-C, no. 10, pp. 1560-1565, 2012.

[32] V. Pierro, V. Galdi, G. Castaldi, I. M. Pinto, and L. B. Felsen,

“Radiation properties of planar antenna arrays based on certain

categories of aperiodic tilings," IEEE Trans. Antennas and Propag.,Vol.

53, No. 2, pp. 635-644, 2005.

[33] Ibrahim, S. M., “Some properties of planar array antennas formed by

symmetry rotation of linear array antenna," 1989 IEEE Antenna and

Propag. Society Int'l Symp. Dig. (AP-S 1989), Vol. 2, pp. 706-708,

90

1989.

[34] E. R. Brown, C. D. Parker, and E. Yablonovitch, “Radiation properties

of a planar antenna onphotonic-crystal substrate," J. Opt. Soc. Am. B,

Vol. 10, No. 2, pp. 404-407, 1993.

[35] I. Toyoda and E. Nishiyama, “Advanced planar antennas integrated

with microwave circuits for RF signal processing applications," Proc.

10th Asia-Pacific Eng. Res. Forum on Microwaves and Electromagnetic

Theory (APMET2014), pp. 111-115, 2014.

[36] H. Sakai, E. Nishiyama, and I. Toyoda, “Direction of arrival estimating

array antenna," Proc. 2012 Int'l Symp. on Antennas and Propag.

(ISAP2012), POS2-24, 2012.

[37] R. Tanaka, E. Nishiyama, and I. Toyoda, “A mono-pulse DOA

estimation antenna integrated with RF amplifiers and detection

circuits," 2014 IEEE Int'l Symp. Antennas and Propag. and USNC-

URSI Radio Sci. Mtg. (2014 AP-S/USNC-URSI) Dig., 526.3, 2014.

[38] R. Rashid, D. Hattori, E. Nisiyama, and I. Toyoda, “An RF multiplier

integrated planar antenna for DOA estimation,” Proc. 2016 Int’l Symp.

on Antennas and Propag. (ISAP2016), POS 1-77, 438-439, Okinawa,

Japan, 2016.

[39] T. Kondo, Y. Ushijima, E. Nishiyama, M. Aikawa, and I. Toyoda,

“Beam steering microstrip array antenna with orthogonal excitation,"

Proc. 2012 Asia-Pacific Microwave Conf. (APMC2012), Vol. 2A5-04,

pp. 67-69, 2012.

[40] K. Sato, K. Nishikawa, and T. Hirako, “Development and field

experiments of phased array antenna for land vehicle satellite

communications," 1992 Antennas and Propag. Society Int'l Symp. (AP-

S 1992) Dig., pp. 1073-1076, 1992.

[41] S. H. Son, S. Y. Eom, and S. I. Jeon, “A novel tracking control

realization of phased array antenna for mobile satellite communication,"

Proc. Vehicular Tech. Conf. 2003 (VTC 2003), pp. 2305-2308, 2003.

[42] R. Rashid, E. Nishiyama, and I. Toyoda, “Prototype evaluation of a

91

beam tracking antenna using magic-T," Proc. 2015 Int'l Symp. on

Antennas and Propag. (ISAP2015), pp. 940-943, 2015.

[43] W. L. Stutzman and G. A. Thiele, “Antenna Theory and Design,” John

Wiley & Sons, New York,1998.

[44] F. C. De Ronde, “A new class of microstrip directional couplers,” 1970

IEEE MTT-S Int'l Microwave Symp. Dig., pp. 184-189, 1970.

[45] H. Wang and D. G. Fang, “A compact single layer monopulse microstrip

antenna array,” IEEE Trans. Antennas and Propag. Vol. 54, No. 2, pp.

503-509, 2006.

[46] W. U. Zhong, G. M. Wang, and C. X. Zhang, “A broadband planar

monopulse antenna array of C band,” IEEE Antenna and Wireless

Propag. Letters, Vol. 8, pp. 1325-1328, 2009.

[47] K. Hemant and G. Kumar, “Microstrip antenna array with ratrace

comparator at X-band for monopulse tracking radar,” 2016 IEEE Int’l

Symp. On Antennas and Propag. (APS-URSI), pp. 513-514, Fajardo,

Puerto Rico, 2016.

[48] Y. Fei, Y. Xie, and L. Zhang, “Single patch antenna with monopulse

patterns,” IEEE Microwave and Wireless Comp. Letters, Vol. 26, pp.

762-764, 2016.

[49] M. A. Hossain, Y. Ushijima, E. Nishiyama, I. Toyoda, and M. Aikawa,

“Orthogonal circular polarization detection patch array antenna using

double balanced RF multiplier,” Prog. in Electromagnetic Research C,

Vol. 30, 65-80, 2012.

[50] L. Liu, and H. Liu, “Joint Estimation of DOA and TDOA of Multiple

Reflections in Mobile Communications,” IEEE Access, Vol. 4, 3815–

3823, 2016.

[51] T. Wang, B. Ai, R. He, and Z. Zhong, “Two-Dimension Direction-of-

Arrival Estimation for Massive MIMO Systems.” IEEE Access, Vol. 3,

2122–2228, 2015.

[52] A. Khabbazibasmenj, A. Hassanien, S. A. Vorobyov, and M. W.

Morency, “Efficient Transmit Beam space Design for Search-Free

92

Based DOA Estimation in MIMO Radar” IEEE Transactions on Signal

Processing, Vol. 62, 1490–1500, 2014.

[53] S. M. Kim, and H. K. Kim, “Direction-of-Arrival Based SNR

Estimation for Dual-Microphone Speech Enhancement,” IEEE/ACM

Transactions on Audio, Speech, and Language Processing, Vol. 22,

2207–2217, 2014.

[54] A. F. Molisch, “Wireless Communications,” ISBN 0-470-84887-1,

Wiley, 2005.

93

Appendix

Antenna Parameters, Dimension

A.1 Magic-T

Figure A-1 Design parameters of Magic-T.

11.1

415

40

2.4

0.2

0.7

Port 1 (50

Port 4

(100

Port 3

(100

Port 2

(200

Unit [mm]

94

A.2 Phase Shifter (Simulation)

Figure A-2 Design parameters of phase shifter.

Port3

Port2Port1

Port4

2.4

1.4

0.7

10.05

11.75

15.5 15.5 9.55

95

A.3 E-plane Beam Tracking Antenna

Figure A-3 Design parameters of E-plane beam tracking antenna.

Port2

Port1

0.2

0.7

2.4

1.4

0.7

11.3

17.16

17.16 0.2

6.25

35.9

11.3

91

45

18.47

127.52

30

0.5

11.75

0.2

10.0513.35

11.75

2.4

96

A.4 H-plane Beam Tracking Antenna

Figure A-4 Design parameters of H-plane beam tracking antenna.

Port 1

Port 2

29

31.85 9.65 25.6

17.12

10.05

66.8

18.35

11.35

90.25

0.7

0.7

2.4

1.4

0.7

22

17.12

11.1

12.35

11.75

0.70.5

97

A.5 DOA Estimation Antenna Integrating RF

Multiplier

Figure A-5 Design parameters of DOA estimation antenna integrating

RF multiplier.

2.4

16.7850.69

45.885

27.45

2.4 6

5.4 2.4

11.28

16.41

3.5

0.2

5.47

5.6

3 1.12

0.4

0.2 10.455

9.65

0.2

8.3

0.69

2

6.36

6

2.65.7

0.20.42.46

24

Unit: mm

3.36

6.37

Port 1

Port 2∑

Δ

55.9

98

A.6 Dual-axis DOA Estimation Antenna

Figure A-6 Design parameters of dual-axis DOA estimation antenna.

Port 4

Port 2

Port 3

yz

xz

yz

Port 1xz

0.1

0.7

0.7

2.4

1.4

0.2

7

35.45

51.25

41.4

7

49.4

85.671.96

2.4

rf signal processing planar antennas for beam tracking and...

Documents