this document is downloaded from cityu …lbms03.cityu.edu.hk/oaps/ee2013-fyp-lsy001.pdfthis...

This document is downloaded from CityU Institutional Repository,

Run Run Shaw Library, City University of Hong Kong.

Title 60 GHz antennas for WiGig/802.11ad

Author(s) Leung, Sut Yee (梁實怡)

Citation Leung, S. Y. (2013). 60 GHz antennas for WiGig/802.11ad (Outstanding Academic Papers by Students (OAPS)). Retrieved from City University of Hong Kong, CityU Institutional Repository.

Issue Date 2013

URL http://hdl.handle.net/2031/7021

Rights This work is protected by copyright. Reproduction or distribution of the work in any format is prohibited without written permission of the copyright owner. Access is unrestricted.

i

Department of Electronic Engineering

FINAL YEAR PROJECT REPORT

BEEECE-2012/13-CHC-06

60 GHz Antennas for WiGig/802.11ad

Student Name: Leung Sut Yee

Student ID:

Supervisor: Prof. C.H. Chan Assessor: Dr. Steve Wong

Bachelor of Engineering (Honours) in Electronic Engineering (Communication Engineering)

ii

Student Final Year Project Declaration

I have read the student handbook and I understand the meaning of academic dishonesty, in particular plagiarism and collusion. I declare that the work submitted for the final year project does not involve academic dishonesty. I give permission for my final year project work to be electronically scanned and if found to involve academic dishonesty, I am aware of the consequences as stated in the Student Handbook. Project Title : 60 GHz Antennas for WiGig/802.11ad

Student Name : Leung Sut Yee

Student ID:

Signature

Date : 22 April 2013

iii

No part of this report may be reproduced, stored in a retrieval system, or transcribed in any form or by any means – electronic, mechanical, photocopying, recording or otherwise – without the prior written permission of City University of Hong Kong.

i

Table of Contents 1 Chapter One Introduction ............................................................................................................... 1

1.1 An overview of IEEE 802.11ad and Wireless Gigabit Alliance ............................................. 1

1.2 Objectives of the research ....................................................................................................... 3

2 Chapter Two Literature review ....................................................................................................... 4

2.1 Literature review ..................................................................................................................... 4

2.2 Significance of the research .................................................................................................... 6

3 Chapter Three Antenna designs and the design rationales .............................................................. 7

3.1 Design 1 ................................................................................................................................... 7

3.1.1 Dimensions revised for 60 GHz antenna ......................................................................... 8

3.1.2 Adjusting the thickness of the upper substrate ................................................................ 9

3.1.3 CPW changed to microstrip line feeding ....................................................................... 10

3.1.4 Impedance matching ...................................................................................................... 12

3.1.5 Simulation results .......................................................................................................... 18

3.1.6 Major cause for the narrowed bandwidth: inductive antenna load ignored ................... 20

3.2 Design 2 (stub for impedance matching) ............................................................................... 21

3.2.1 Matching the inductive antenna load by adding a microstrip stub ................................ 21


3.2.3 Problems in the radiation patterns ................................................................................. 24

3.3 Design 3 (three-layer structure) ............................................................................................. 25

3.3.1 Major revisions in the structure ..................................................................................... 26

3.3.2 A middle substrate of 0.127 mm and placing the transmission line in the middle ........ 27

3.3.3 Increasing the height of the VPA .................................................................................. 28

3.3.4 Air cavity wall coated with copper ................................................................................ 28

3.3.5 A completed model after revision ................................................................................. 29


3.3.7 Measurement results ...................................................................................................... 33

3.3.8 Possible reasons for failure – air gap between the upper and the middle substrate ....... 34

3.4 Design 4 (Design 1 with enlarged air cavity) ........................................................................ 35

3.4.1 Simulation results of the completed model with soldering pin ..................................... 36

ii

3.4.2 Comparing measurement and simulation results ........................................................... 40

3.4.3 Results summarized ....................................................................................................... 53

4 Chapter Four Discussions ............................................................................................................. 55

4.1 An attempt to develop a theoretical VPA model ................................................................... 55

4.2 Explaining the discrepancies between measured and simulated resonances and bandwidths59

4.3 Satisfying the 802.11 ad central frequency and bandwidth requirements ............................. 59

4.4 Radiation patterns explained ................................................................................................. 61

4.4.1 E-plane Co-polar ........................................................................................................... 62

4.4.2 H-plane Co-polar ........................................................................................................... 64

4.4.1 E-plane Cross-polar (explaining the large measured E-X) ............................................ 65

4.4.1 H-plane Cross-polar (explaining the large measured H-X) ........................................... 66

4.5 Large null at �~��° on E-Co ............................................................................................. 67

4.6 Orientation of the antenna in application .............................................................................. 68

4.6.1 E-plane: received power and vertical beamwidth .......................................................... 68

4.6.2 H-plane: received power and horizontal beamwidth ..................................................... 69

4.7 Beamwidth for WLAN applications ...................................................................................... 70

4.8 Difference between co-polarization and cross-polarization .................................................. 71

4.9 Antenna performances at other frequencies .......................................................................... 71

4.10 Gain ....................................................................................................................................... 73

4.11 Air cavity (explaining how gain is amplified) ....................................................................... 74

5 Chapter Five Further research ...................................................................................................... 76

5.1 Feeding design 1 (CPW) ................................................................................................... 76

5.2 Feeding design 2 (CPW with smaller top ground) ............................................................ 79

5.3 Feeding design 3 (Probe feed) ........................................................................................... 81

5.4 Feeding design 4 (Probe feed with transmission line) ....................................................... 84

5.5 A framework for further research ...................................................................................... 87

6 Chapter Six Conclusions .............................................................................................................. 88

References ............................................................................................................................................. 89

iii

List of figures Figure 1 : Antenna by Wong, Ng, Luk, Chan and Xue (2010) – side view ............................................ 7

Figure 2: Antenna by Wong, Ng, Luk, Chan and Xue (2010) – orthogonal view .................................. 8

Figure 3: Reflection coefficient for ��() = . �� .................................................................. 10

Figure 4: Reflection coefficient for h1(60) = 0.254mm .................................................................... 10

Figure 5: Side view of Design 1 (dielectric constant of Duroid 5880 �� = �. �) ................................. 11

Figure 6: Top view of Design 1 ............................................................................................................. 12

Figure 7: Impedance matching for Design 1 ......................................................................................... 13

Figure 8: Impedance matching equivalent circuit for Design 1 ............................................................ 13

Figure 9: Antenna impedance with excitation directly applied to the antenna...................................... 15

Figure 10: Widest possible bandwidth by perfect matching ................................................................. 16

Figure 11: Design 1 reflection coefficient ............................................................................................. 19

Figure 12: Design 1 Smith chart ............................................................................................................ 20

Figure 13: Design 2 top view ................................................................................................................ 22

Figure 14: Design 2 equivalent circuit .................................................................................................. 23

Figure 15 Smith Chart for stub width set to 0.198mm .......................................................................... 23

Figure 16: Reflection coefficient for stub width set to 0.198mm .......................................................... 24

Figure 17: Radiation patterns for stub width set to 0.198mm ............................................................... 25

Figure 18: Side view of Design 3 .......................................................................................................... 26

Figure 19: E-fields directed to ground .................................................................................................. 27

Figure 20: increasing the height of VPA ............................................................................................... 28

Figure 21: No copper coating ................................................................................................................ 29

Figure 22: With copper coating ............................................................................................................. 29

Figure 23: Design 4 complete model side view .................................................................................... 30

Figure 24: Reflection coefficient of Design 3 (simulation) ................................................................... 30

Figure 25: Radiation pattern of Design 3 (simulation) .......................................................................... 31

Figure 26: Gain of Design 3 (simulation).............................................................................................. 32

Figure 27: Fabricated antenna of Design 3............................................................................................ 33

Figure 28: Return loss of Design 3 (measurement) ............................................................................... 33

Figure 29: Air gap between upper and lower substrate ......................................................................... 34

Figure 30: Return loss of Design 4 (simulation) ................................................................................... 36

Figure 31: A completed model of Design 4 .......................................................................................... 37

Figure 32: Return loss of Design 4 (simulation) ................................................................................... 38

Figure 33: Radiation patterns of Design 4 (simulation) ........................................................................ 39

Figure 34: Gain of Design 4 (simulation).............................................................................................. 40

Figure 35: Fabricated antenna of Design 4............................................................................................ 40

Figure 36: Return loss of Design 4 (measurement and simulation) ...................................................... 41

Figure 37: VSWR of Design 4 (measurement and simulation) ............................................................. 42

iv

Figure 38: Radiation patterns at 60 GHz of Design 4 (measurement) .................................................. 43

Figure 39: Radiation patterns at 60 GHz of Design 4 (simulation) ....................................................... 44

Figure 40: E-Co at 60 GHz of Design 4 (measurement and simulation) .............................................. 45

Figure 41: E-X at 60 GHz of Design 4 (measurement and simulation) ................................................ 46

Figure 42: H-co at 60 GHz of Design 4 (measurement and simulation) ............................................... 47

Figure 43: H-X at 60 GHz of Design 4 (measurement and simulation) ................................................ 48





Figure 48: Gain of Design 4 (measurement and simulation)................................................................. 53

Figure 49: Explanation for the unbalanced E-X .................................................................................... 54

Figure 50: Radiation of a patch antenna ................................................................................................ 55

Figure 51: Annular patch antenna developed to circular VPA .............................................................. 56

Figure 52: 60 GHz band spectrum in different countries ...................................................................... 60

Figure 53: Radiation patterns defined ................................................................................................... 61

Figure 54: Bessel function of first kind ................................................................................................. 62

Figure 55: E-plane Co-polarization ....................................................................................................... 63

Figure 56: H-plane Co-polarization ...................................................................................................... 64

Figure 57: E-plane Cross-polarization .................................................................................................. 65

Figure 58: H-plane Cross-polarization .................................................................................................. 66

Figure 59: E-plane polarization of the VPA at 33 GHz ........................................................................ 67

Figure 60: Radiation caused by an SMA connector .............................................................................. 68

Figure 61: Received signal on E-plane .................................................................................................. 69

Figure 62: Received signal on H-plane ................................................................................................. 70

Figure 63: Standing waves .................................................................................................................... 74

Figure 64: Reflected E-fields from the air cavity .................................................................................. 75

Figure 65: E-field directed to the top ground ........................................................................................ 76

Figure 66: Top view of Feeding design 1 .............................................................................................. 77

Figure 67: Reflection coefficient of Feeding design 1 .......................................................................... 78

Figure 68: Explanation for the ripples ................................................................................................... 78

Figure 69: Top view of Feeding design 2 .............................................................................................. 79


Figure 71: Radiation patterns of Feeding design 2 ................................................................................ 81

Figure 72: Feeding design 3 – probe feed ............................................................................................. 82



Figure 75: Feeding design 4 – probe feed with transmission line ......................................................... 84



v

List of tables

Table 1: Different generations of 802.11 ................................................................................................ 1

Table 2: Normalized gains at 60 GHz ................................................................................................... 43



Table 5: Zeros for Bessel function of first kind .................................................................................... 57

Table 6: Eigenvalues for circular modes ............................................................................................... 57

Table 7: Effective radius at different frequencies ................................................................................. 58

Table 8: Antenna performance at 55 GHz ............................................................................................. 71

Table 9: Antenna performance at 60 GHz ............................................................................................. 71

Table 10: Antenna performance at 65 GHz ........................................................................................... 72

Table 11: Gains of other antennas ......................................................................................................... 73

vi

Abstract

This project is an attempt to develop an antenna with a center frequency at 60 GHz and a

bandwidth of 7 GHz for the WiGig 802.11ad applications. The proposed antenna is a circular

vertical patch antenna (VPA) with two layers of Duroid 5880 as substrates. An air cavity is

embedded in the lower substrate for gain enhancement. The antenna is fed by microstrip

transmission line placed on the top of the upper substrate. Measurement results show the

antenna obtains an impedance bandwidth of 15 GHz (25%) from 51.5 GHz to 66.5 GHz; and

yields a gain of 8.3 dBi at 60 GHz. Radiation patterns at 55 GHz, 60 GHz and 65 GHz were

measured. The antenna satisfies the requirements for the 802.11 ad applications.

1

1 Chapter One Introduction

1.1 An overview of IEEE 802.11ad and Wireless Gigabit Alliance

802.11 is a set of wireless communication standards set by IEEE. The standard was first

introduced in 1997. Over the years, the standards have been revised for a number of times

with additional features to address the increasing demand for wireless communication. The

first version of 802.11 only supports web browsing. With revised version in 2003, the needs

for data and file sharing are catered. As technology progresses, markets are now looking for

video and audio signals communication by wireless means. The 802.11ac and 802.11ad are

announced in this regard. Brief descriptions for different 802.11 standards are listed as

follows:

IEEE

standard Data rate Coverage Wireless applications

802.11 a 54 Mbps ~ 35 m Web browsing

802.11 b 11 Mbps ~ 78 m Web browsing

802.11 g 54 Mbps ~ 78 m Data and file sharing

802.11 n 600 Mbps ~ 75 m Sending/receiving data and video signals

802.11 ac 3.2 Gbps ~ 30 m Sending/receiving audio and video signals

802.11 ad 7 Gbps < 10 m Sending/receiving uncompressed audio and video

signals

Table 1: Different generations of 802.111 2

1 CTimes http://www.ctimes.com.tw/DispNews-tw.asp?O=HJX1H6UT0FKSAA0MEC 2 S. Sendra, M. Garcia, C. Turro, and J. Lloret, “WLAN IEEE 802.11a/b/g/n indoor coverage and interference

performance study”, International journal on advances in networks and services, vol. 4, No 1&2, pp.209-222, 2011.

2

To be more specific, 802.11 ad is the latest wireless communication standard with a central

frequency at 60 GHz and a data rate of 7Gbps. The high data transmission rate allows

sending and receiving audio and video signals by wireless channels, thus saving the cost of

using HDMI cables. The 60 GHz band is so chosen to avoid the interference with lower band

standards such as Wi-Fi. It is also an unlicensed band in most of the countries.

Wireless Gigabit Alliance (WiGig) is an organization promoting the IEEE 802.11 ad standard

among consumer electronics such as cell phones, handheld devices, PCs, televisions and

cameras. With the purpose of establishing a universal environment for 60GHz 7Gbps

wireless communication, it specifies the hardware architectures, packet structures and

communication protocols which satisfy the 802.11 ad standard. Products adopting the WiGig

specifications are interoperable with each other. Several major features of the WiGig

specification are highlighted:

• Data transmission rate up to 7 Gbps

• Compatibility with lower band standard such as Wi-Fi at 2.4 GHz and 5 GHz

• Beamforming allowing communication beyond 10 meters

• Adopting the Advanced Encryption Standard (AES) to enhance the security in

wireless communication

• Power saving by scheduled communication periods which is important for battery-

powered devices

The 802.11ad standard, however, does have its limitation. The free space propagation loss

over 1m at 60 GHz is 68dB as compared to 38.4dB at 5 GHz with the same input power.

Moreover the RF absorption peak at 60 GHz becomes noticeable for distance beyond 100m.

This limits the maximum distance for wireless communication.3

With its features of short distance, high data rate communication, WiGig will find its

applications in WLAN and WPAN. In particular, WiGig is suitable for wireless high 3 “Wireless Lan at 60 GHz – IEEE 802.11ad Explained” by Agilent Technologies”

http://cp.literature.agilent.com/litweb/pdf/5990-9697EN.pdf

3

definition audio/video signals transmission within indoor environment. Examples include the

wireless connections between PCs, projectors, monitors, digital cameras, digital video

recorders and televisions.

The market penetration of WiGig products is currently low as the finalized version of

802.11ad was just published in the beginning of 2013. Nevertheless it is expected that a

wealth of WiGig products will be available by the end of 2014.

1.2 Objectives of the research

In view of the upcoming trend in the WiGig market, this research is an effort devoted to

develop an antenna which is in accord with the 802.11ad standard. In particular the antenna

should satisfy the following requirements:

• Central frequency of 60GHz

• Absolute bandwidth of 7 GHz to support the data rate of 7Gbps

• Compact in size and low fabrication cost in order to be widely adopted in consumer

electronics

This report will begin by reviewing the literature on related research. The major part of this

report will be the details on how different antenna models are developed; the design

rationales, the simulation and the measurement results will be included in this section. In the

discussion pages, the antenna operation mechanism, measurement results and the antenna

applications will be thoroughly explained. Another section will be devoted to outline further

trials to improve the antenna performance. The report will then end by evaluating the extent

to which the antenna satisfies the 802.11 ad requirements.

4

2 Chapter Two Literature review

2.1 Literature review

Facing the surging demand for wireless gigabit data transmission, various types of high

frequency antennas have been developed over the years for this purpose. Liu, Akkermans and

Floyd (2009) presented an aperture-coupled patch antenna which can be integrated with RFIC.

The measured gain and bandwidth are 7dBi and 12 GHz respectively with the air cavity.

Zhang and Sun (2009) reported a design of grid array antenna which avoids the problem of

complex feeding method. The 60 GHz antenna is made up of rectangular meshes of

microstrip lines on a dielectric substrate fed by metal via through aperture on the ground to

attain a bandwidth of13 GHz and peak realized gain of 15 dBi.

Sun et al. (2008) designed a Yagi antenna with five ceramic layers of which four are cavity

layers to hold a radio chip. The second layer holds the Yagi driver and four Yagi directors

and the third and fifth layers hold the reflector. The resulting bandwidth is 2.3GHz from 60.6

to 62.9 GHz and a peak gain of 6dBi.

Grzyb et al (2006) printed the 60 GHz antenna on the bottom of a substrate which is

suspended over the reflected ground. The substrate is surrounded by a metal frame for

curbing the loss between the substrate and the ground and thus ensuring the wide bandwidth.

The bandwidth and the peak gain are 12 GHz and 8.5 dB respectively. The antenna appears to

be insensitive to the surrounding package, PCB dielectric and cavity manufacturing.

In the design of Kim et al (2005), a radiating patch which functions as an antenna is

supported by two posts and fed by a CPW and a feeding post. The bandwidth of 5.8 GHz

from 58.7 to 64.5 GHz is achieved. This antenna is a system-on-chip (SOC) and can be

integrated with monolithic microwave integrated circuit (MMIC).

A four-element microstrip patch antenna was proposed by Park and Wang (2003). The design

is integrated with in-phase/quadrature-phase mixer to process high-speed signal at 60 GHz.

Bandwidth about 2 GHz is obtained by this antenna.

5

Lin et al (2007) fabricated a triangular monopole antenna fed by CPW. This is a CMOS

RFIC-on-chip antenna. The bandwidth so obtained is 8 GHz and the peak gain is –15 dB.

This antenna is also isotropic.

Two 60 GHz antennas were presented by Zhang, Sun and Guo (2005). One is an inverted-F

antenna and the other one quasi-Yagi antenna. These designs are novel in the sense that the

inverted-F and quasi-Yagi antenna are printed by a low resistivity silicon substrates. The

inverted-F antenna demonstrates a bandwidth of 10 GHz but the lowest return-loss for the

quasi-Yagi antenna is only –6.75 dB. The gains at resonance frequency are –19 dBi and –12.5

dBi for the inverted-F and the Yagi antenna respectively.

Gutierrez et al (2009) designed three antennas at 60 GHz for WPAN applications. These are

half-wave dipole, Yagi and rhombic antennas. Bandwidth for both dipole and Yagi antenna

are approximately 12 GHz while that for the rhombic antenna is 7 GHz. However the

rhombic antenna demonstrates the highest peak gain of –0.2 dBi and that for the dipole and

Yagi antenna are –7.3 dBi and –3.55 dBi respectively.

Sironen, Qian and Itoh (2001) proposed a conical horn antenna to attain a 16.5dBi gain at

60GHz and a -11dB return-loss over 4 GHz.

Another 60GHz high gain antenna was developed by Karnfelt et al (2006) with the structure

of a microstrip array which was fabricated on the same substrate with an 18dB gain amplifier.

Pavuluri, Fersidis, Goussetis and Wang (2008) adopted a suspended frequency selective

surface (FSS) in their antenna design. With the FSS based cavity, a directivity of 25dBi was

achieved at 60GHz.

A brief review of the recently developed antennas finds that most of them are in conventional

designs such as dipole, Yagi, array, monopole horn, patch and inverted-F. In fact, there is an

impending trend in vertical patch antenna (VPA). Mak, Luk and Lee (2003) presented a VPA

with the advantages of being more compact in size and improved wideband. Experimental

results showed a gain of 8dBi and over 10% bandwidth at 5.6GHz.

6

The above mentioned VPA was then modified for dual-band operation by Lau, Wong, Mak,

Luk and Lee (2006). The resulting bandwidths were 7% and 26% at 4.25GHz and 7GHz

respectively while both bands achieved a gain of 7dBi.

This VPA was further developed for millimeter applications by Wong, Ng, Luk, Chan and

Xue (2010). This new type of antenna has an air cavity as an additional feature for gain

improvement and a grounded CPW in place of the probe-feed. A gain of 7.9dBi and 18%

bandwidth were attained with 33GHz as the resonance frequency. This antenna is fabricated

using PCB and plated-through-hole technologies, which are common technologies and incur

low cost.

2.2 Significance of the research

There are already a number of 60 GHz antennas. However some of them either attain low

gain or have narrow bandwidth. Although others can achieve high gain and wide bandwidth,

their structures are complex or large in size, which implies a high manufacturing cost. VPA,

on the other hand, can be fabricated using low cost technology though at a lower band, say 33

GHz. This research works towards the application of low cost technology to the 60 GHz

antenna. This would be a novel attempt to develop a 60 GHz antenna in VPA. The antenna

so developed can satisfy the WiGig requirements in an economical way.

7

3 Chapter Three Antenna designs and the design rationales

This section gives the details on how and why different antenna models are built. Latter

models are developed based on the failure analysis of the previous ones.

3.1 Design 1

The first design is based on the antenna presented in the paper “Vertical patch antenna for

millimeter wave applications” by Wong, Ng, Luk, Chan and Xue (2010) for the following reasons:

• It is a VPA, the structure which this research aims to develop

• Its central frequency is 33 GHz, which is the highest frequency at which VPA has been

applied to

• It is simple in structure – planar, substrate layers and no complex feeding network, which

ensures the low fabrication cost

The structure as presented in the paper is shown in the following diagram.

Figure 1 : Antenna by Wong, Ng, Luk, Chan and Xue (2010) – side view

8

Figure 2: Antenna by Wong, Ng, Luk, Chan and Xue (2010) – orthogonal view

The dimensions are also listed for reference: D� = 2.4 mm (0.264λ) D� = 8 mm h� = h� = 0.787 mm

r = 0.18 mm (0.019λ) L = W = 18 mm (2λ) Duroid 5880 is used for S�, S�

3.1.1 Dimensions revised for 60 GHz antenna

The size of an antenna is inversely proportional to its central frequency. The 33 GHz antenna

needs to be scaled down for the 60 GHz application. The values of D�, r, L and W are given

as a proportion of wavelength. These fractions are used to find the dimensions of the

corresponding dimensions of the 60 GHz antenna. Calculations are outlined as follows:

9

AT 60 GHz wavelength,

λ"# = $60 × 10& = 5((

D� = 0.264λ = 1.32mm r = 0.019λ = 0.095mm

This value is too small for HSFF simulation, so it is changed to 0.099 mm. L = W = 2λ = 10mm

The dimension of D�, h� and h� are scaled proportionally to wavelength. D�("#)λ"# = D�(--)λ--

D�("#) = 4.4((

h�("#)λ"# = h�(--)λ--

h�("#) = h�("#) = 0.4328((

A substrate of thickness 0.4328 mm is not available for fabrication, so h�("#) and h�("#) are

changed to 0.508 mm.

3.1.2 Adjusting the thickness of the upper substrate

However if h�("#) is set to be 0.508 mm, no resonance is observed around 60 GHz and if

h�("#) is reduced to 0.254 mm, simulation result shows a resonance at 60 GHz. Therefore the

thickness of the upper substrate is chosen to be 0.254mm. Reflection coefficients for h�("#) =0.508(( and h�("#) = 0.254(( are shown in the following two graphs.

10

Figure 3: Reflection coefficient for ��() = . ��

Figure 4: Reflection coefficient for h�("#) = 0.254mm

3.1.3 CPW changed to microstrip line feeding

Besides the revisions in dimensions, consideration has also been given to the feed line. The

33 GHz antenna is fed by CPW placed between the two substrates. Difficulties arise when the

11

CPW is connected to the connector. The edge of the substrates has to be cut for the

connection. The damage made to the substrates may adversely affect the antenna

performance. Therefore in the revised design, the transmission line is placed on the top as the

feed line.

As a whole, the revised structure and dimensions are as follows:

Figure 5: Side view of Design 1 (dielectric constant of Duroid 5880 �� = �. �)

12

Figure 6: Top view of Design 1

To summarize, the revised antenna has the following characteristics:

� There are two substrates of Duroid 5880 /0 = 2.2

� The VPA is inserted in the upper substrate

� A ground plane is placed between the upper substrate and the lower substrate

� A reflected ground is under the lower substrate

� An air cavity is inserted in the lower substrate

� The VPA is fed by a transmission line, which is placed on the top of the upper substrate.

3.1.4 Impedance matching

The antenna will be excited by EM wave delivered by a connector of 50 1. It is necessary for

the impedance of the antenna matched to that of the connector in order to maximize the

13

power transferred to the input port of the antenna. Quarter wave transformation technique

will be used. The following paragraph explains how the matching is done with reference to

the following layout and equivalent circuit.

Figure 7: Impedance matching for Design 1

Figure 8: Impedance matching equivalent circuit for Design 1

14

Finding 2 and 2′

Input impedance 456 = 501

Transmission line impedance 4# = 501

Quarter wave transformer impedance 4#7 Antenna impedance 48

Length and width of the transmission line are 2 and 9 respectively.

Length and width of the quarter wave transformer are 2′ and 9′ respectively.

27 = λ"#(:;;)4 = 5((4√2.2 = 0.8427((

2 = >2 − D�2 − 27 = 5(( − 2.2(( − 0.8427(( = 1.9573((

48, 4#7 , 9 and 9′ are to be found so that 48 can be matched to the input impedance of 50 1.

Finding 48

To find 48, the quarter wave transformer and the transmission line are ignored. Excitation is

directly applied to the antenna by a 50 1 source. The impedance at 60 GHz is obtained by

simulation, which is taken as 48 . The simulation results are presented in the following

diagram.

15

Figure 9: Antenna impedance with excitation directly applied to the antenna

From the simulation result, 48 is 97.6 + A18.771.

In order to find the widest possible bandwidth, the antenna is excited directly by a source

with a hypothetical impedance 456′ = 97.6 − A18.771, which is a conjugate of 456. This is to

simulate the situation of perfect matching. The widest possible bandwidth is found to be 11.5

GHz shown as below.

16

Figure 10: Widest possible bandwidth by perfect matching

Finding 4#7 By quarter wave transformation, 4#7 can be found using the following formula.

456 = 4#7 �48

50 = 4#7 �97.6 + A18.771

4#7 = 70.17 + A6.661

Both real and imaginary part exist in 4#7 . Unfortunately quarter wave transformation is

limited to matching real impedance. However the imaginary part is only 9% of the real part.

Therefore it is ignored in the matching.

Finding 9 and 9′

It is necessary to find the width and both the transformer and the transmission line so that

their impedance are 70.17 1 and 50 1 respectively.

17

The following empirical formula relating width of a microstrip line and its impedance are

used.

9:;;ℎ = 8CDC�D − 29:;;ℎ < 2

9:;;ℎ = 2F GH − 1 − ln(2H − 1) + /0 − 12/0 Kln(H − 1) + 0.39 − 0.61/0 LM9:;;ℎ > 2

where

O = 4# 2FP# Q/0 + 12 + /0 − 1/0 + 1R0.23 + 0.11/0 S

H = 14# FP#2√/0 9:;; = 9TUV − 1.44ℎ

9:;; – effective width of a microstrip line

9TUV – physical width of a microstrip line

Let 9:;; and 9:;;7 be the effectively width of the transmission line and the transformer

respectively. Using the formula, 9:;;ℎ� = 3.1263

9:;;7ℎ� = 1.7901

9:;; = 1.5882mm

9:;;7 = 0.9094((

9 = 2.3197mm 97 = 1.6409((

18

Completed set of parameters

Input port: 456 = 501

Transmission line 4# = 501 2 = 1.9573(( 9 = 2.3197mm

Transformer 4#7 = 70.171 27 = 0.8427(( 97 = 1.6409((

3.1.5 Simulation results

The reflection coefficient is mainly referred to as a preliminary study to see if an impedance

bandwidth of 7 GHz can be attained. With the above complete set of parameters, reflection

coefficient is found as follows.

19

Figure 11: Design 1 reflection coefficient

It is obvious from the reflection coefficient results that a bandwidth of 7 GHz cannot be

attained. Compared this with the widest possible bandwidth of 11.5 GHz, the most possible

reason for the narrowed bandwidth is that the antenna impedance is fail to match with the

source impedance.

20

3.1.6 Major cause for the narrowed bandwidth: inductive antenna load ignored

To investigate the problem, Smith chart is referred.

Figure 12: Design 1 Smith chart

A locus of normalized antenna impedance from 55 GHz to 70 GHz is shown in the Smith

chart. It is known from the Smith chart that the antenna impedance is inductive as it is on the

upper plane of the chart, which is consistent with the result of 48 = 97.6 + A18.771 presented

before. In order to attain a wider bandwidth, say from 55 GHz to 65 GHz as in the case of perfect

21

match, the corresponding locus must lie within the circle for VSWR = 1.925, which corresponds to a W�� value of −10XH according to the following calculation.

W��(XH) = −20 log W�� = −10XH W�� = 3.1623 [8 = W�� = 3.1623

VSWR = [8 + 1[8 − 1 = 3.1623 + 13.1623 − 1 = 1.925

The 1.925 VSWR circle is marked on the above Smith Chart. Due to the inductive nature of the

antenna load, a section of the locus from 55 GHz to 65 GHz is out of the 1.925 VSWR circle.

Therefore it is suspected that excluding the imaginary part of the antenna impedance is considered to

be the main reason for such a mismatch and narrowed bandwidth.

3.2 Design 2 (stub for impedance matching)

3.2.1 Matching the inductive antenna load by adding a microstrip stub

The above failure results suggest that the inductive nature of the antenna load must be taken

into account for impedance matching. A capacitor should be added to the transmission line

for matching the inductive load. The capacitor is realized by an open end microstrip stub. The

following equations show how a capacitive load is realized by an open end stub.

X_ = 1A`a

Z56 = 4#" 48 + A4#" tan f24#" + A48 tan f2 48 → ∞ = 4#"A tanf2 2 = i8

= 4#"A = 1A`a jk4#" = 1̀a

22


It is shown that a capacitor can be implemented with open end microstrip stub if a suitable

value of characteristic impedance is chosen. However the capacitance of the stub required to

matching the inductive load so that the locus from 55 GHz to 65 GHz can lie within the 1.925

VSWR is unknown. So it is by a trial-and-error approach to find the width of the stub for

inductive load matching. The top view of the design and the equivalent circuit are shown in

the following figures.

Figure 13: Design 2 top view

23

Figure 14: Design 2 equivalent circuit

0.198 mm is an arbitrary value set to be the stub width to study the effect of adding an open

end stub. The Smith Chart and return-loss are shown below.

Figure 15 Smith Chart for stub width set to 0.198mm

24

Figure 16: Reflection coefficient for stub width set to 0.198mm

According to the Smith Chart, a stub with width 0.198 mm matches the inductive load of the

antenna and the impedance locus from 55 GHz to 70 GHz now lies within the 1.925 VSWR.

This result agrees with the reflection coefficient figure as the antenna has a wide bandwidth

of about 15GHz.

3.2.3 Problems in the radiation patterns

Although a wide bandwidth can be achieved with an open end stub, another problem arises

with the radiation patterns. Refer to the following radiation patterns, the E-Co pattern is

distorted. The gain is only –4.647 dB at 30°. Moreover the H-X is too large. The H-X at 46° has a gain of 5.074 dB, which is greater than the H-Co. This distorted radiation patterns may

be a result of adding a stub. This extra perfect conductor on the top of the substrate increases

the radiation loss.

25

Figure 17: Radiation patterns for stub width set to 0.198mm

3.3 Design 3 (three-layer structure)

In fact the radiation problem mentioned above does not only exit in the stub. As the

transmission line is put on the top of the upper substrate and it is exposed to the air, part of

the radiated power is radiated to free space along the whole transmission line instead of fed to

the VPA.

26

3.3.1 Major revisions in the structure

High radiation loss is a drawback of microstrip line. The radiation patterns are therefore

inevitably distorted and high gain can hardly be achieved for the range −30° l θ l 30° . To

curb the radiation loss, the antenna structure is revised. The following changes are made:

1. A substrate Duroid 5880 of thickness 0.127 mm is inserted between the upper and lower

substrates.

2. The transmission line is placed between the middle and upper substrates instead of on

the top.

3. Thickness of the upper substrate and the height of the VPA increase to 0.508mm.

4. The air cavity wall in the lower substrate is coated with copper.

Figure 18: Side view of Design 3

The following paragraphs explain the reasons for such changes.

27

3.3.2 A middle substrate of 0.127 mm and placing the transmission line in the middle

The aim of these changes is mainly to suppress the radiation loss. In an ideal antenna, the

transmission line should be close to the ground so that the electric field can be directed to the

ground instead of being radiated. On the other hand, the resonance structure should be placed

as high as possible for inducing stronger radiation intensity. However it is not possible to

include both characteristics in previous design. It is because increasing the thickness of the

upper substrate increases the VPA height at expenses of extended distance between the

transmission line and ground. The result is that a high gain cannot be achieved.

To correct this problem, a middle substrate is inserted and the transmission line is placed

between the upper and the middle substrates. The middle substrate should be as thin as

possible such that the transmission line can be close to the ground so that E-fields are directed

to the ground as shown in the following diagram. With this revised structure, adjusting the

VPA height is independent of the transmission line-ground distance.

Figure 19: E-fields directed to ground

28

3.3.3 Increasing the height of the VPA

Continue with previous sub-session, the purpose of increasing the height of the VPA is to

enhance the radiation intensity to achieve high gain. As the height of the VPA increases,

there is larger conducting surface to allow more moving free charges. This implies that more

E-fields are induced, which contribute to the radiation.

Figure 20: increasing the height of VPA

3.3.4 Air cavity wall coated with copper

The objective of coating the wall of the air cavity is to increase the gain. With the coated wall,

more incident waves are reflected.

The following figures compare the radiation with and without the copper coating. Without the

copper coating, a certain proportion of incident waves on the air cavity wall transmits from

free space into the dielectric and only part of the incident waves are reflected. However with

the copper coating which is a conducting plane, most of the incident waves are reflected. The

reflected waves superimpose with the radiated waves and results in higher gain.

29

Figure 21: No copper coating

Figure 22: With copper coating

3.3.5 A completed model after revision

To study the performance of the antenna under the real experiment set up, a completed model

is built for simulation. The complete model resembles the system that will be set up for

testing the antenna. In the complete model, the following features are added:

1. A base on which the antenna will be placed for testing

2. Four screws which will be used to fixed the antenna on the base

3. A copper layer placed between the antenna and the base

30

Figure 23: Design 4 complete model side view


Figure 24: Reflection coefficient of Design 3 (simulation)

31

Figure 25: Radiation pattern of Design 3 (simulation)

32

Figure 26: Gain of Design 3 (simulation)

As noted from the simulation results, bandwidth covers the range from 49 GHz to 66 GHz

and an average gain is more than 7 dBi within the range of interests although the E-Co

pattern is still asymmetric in its shape. However as this antenna design already satisfies the

central frequency, bandwidth and gain requirements, it will be fabricated for measurement.

33

3.3.7 Measurement results

The fabricated antenna of Design 3 is shown below.

Figure 27: Fabricated antenna of Design 3

Figure 28: Return loss of Design 3 (measurement)

34

It is shown in the above figure that the fabricated antenna fails to satisfy the return loss

requirement. The lowest return loss attained is –8.25 dB which is higher than the benchmark

of is –10 dB. As the antenna fails to satisfy the return loss requirement, it is not necessary to

take further measurements on radiation pattern and gain. Instead the causes for this failure

should be investigated.

3.3.8 Possible reasons for failure – air gap between the upper and the middle substrate

To connect the antenna to the measurement equipment such as a vector network analyzer, a

pin is soldered on the transmission line which is then inserted to an SMA connector. This

results in an air gap appeared between the upper and the lower substrates after the pin is

soldered. The pin is about 0.3mm in diameter. A reasonable estimation for the height of the

air gap would approximately be 0.3mm. The effective wavelength of 60 GHz in Duroid 5880

(/0 = 2.2) is 3.77mm. This means the air gap is comparable to 0.1λ.. This air gap is not

included in the simulation model. This leads to the conclusion that the air gap attributes to the

significant radiation loss, so the measured results deviates from the simulated one.

Figure 29: Air gap between upper and lower substrate

35

3.4 Design 4 (Design 1 with enlarged air cavity)

Due to the previous failure, there is a revision in the antenna design. As placing the

microstrip line between the upper and middle substrate inevitably lead to the air gap, it is

necessary to consider Design 1 again where the microstrip line is put on the top of the

substrate and there are only two substrates instead of three. However the major problem with

Design 1 is the narrow bandwidth. So it is important to devise other methods to enhance the

bandwidth of Design 1.

One way to enhance the bandwidth is to increase the radius of the air cavity. The air cavity

radius of Design 4 is increased from 2.2 mm (Design 1) to 2.37mm. It is because the diameter

of the air cavity and the wavelength should be confined to the relation 2n = oλ, where D is

the air cavity diameter. The bandwidth increases to about 12 GHz with the increased air

cavity dimension.

36

Figure 30: Return loss of Design 4 (simulation)

3.4.1 Simulation results of the completed model with soldering pin

Similar to Design 3, a completed model for Design 4 is built to simulate the antenna

performance under realistic measurement condition. In this time, a soldering pin, a connector

and the 50 ohm coaxial cable are also included to the model. Simulation results are shown in

the following figures.

37

Figure 31: A completed model of Design 4

38

Figure 32: Return loss of Design 4 (simulation)

39

Figure 33: Radiation patterns of Design 4 (simulation)

40

Figure 34: Gain of Design 4 (simulation)

The antenna is fabricated after checking the bandwidth and central frequency requirements

with the simulated results.

3.4.2 Comparing measurement and simulation results

Figure 35: Fabricated antenna of Design 4

41

The ring radius of the fabricated antenna is 0.65 mm instead of 0.62 mm in the design due to

the limitation in fabrication technology. The measurement results as shown in the folllowing

figures are compared with the simulation results based on 0.65 mm ring radius. (The

simulation results shown above are based on the model with 0.62 mm ring radius)

Reflection coefficient and VSWR

Figure 36: Return loss of Design 4 (measurement and simulation)

42

Figure 37: VSWR of Design 4 (measurement and simulation)

0

0.5

1

1.5

2

2.5

50 55 60 65 70

VS

WR

Frequency (GHz)

Measured VSWR

Simulated VSWR

43

Radiation patterns

Figure 38: Radiation patterns at 60 GHz of Design 4 (measurement)

Normalized gain (dB) by measurement p = 330° p = 0° p = 30° E-Co −1.801 −2.625 −14.625 E-X −36.964 −21.412 −35.527 H-Co −2.453 −0.156 −1.568 H-X −29.672 −19.745 −27.527 Table 2: Normalized gains at 60 GHz

-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330

Measured patterns (60 GHz)

Measured E-CoMeasured E-XMeasured H-CoMeasured H-X

44

Figure 39: Radiation patterns at 60 GHz of Design 4 (simulation)

-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330

Simulated patterns (60 GHz)

Simulated E-CoSimulated E-XSimulated H-CoSimulated H-X

45

Figure 40: E-Co at 60 GHz of Design 4 (measurement and simulation)

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150

180

210

240

270

300

330

Measured and simulated E-Co (60 GHz)

Measured E-Co

Simulated E-Co

46

Figure 41: E-X at 60 GHz of Design 4 (measurement and simulation)

-60

-50

-40

-30

-20

-10

0

100

30

60

90

120

150

180

210

240

270

300

330

Measured and simulated E-X (60 GHz)

Measured E-X

Simulated E-X

47

Figure 42: H-co at 60 GHz of Design 4 (measurement and simulation)

-40

-35

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150

180

210

240

270

300

330

Measured and simulated H-Co (60 GHz)

Measured H-Co

Simulated H-Co

48

Figure 43: H-X at 60 GHz of Design 4 (measurement and simulation)

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150

180

210

240

270

300

330

Measured and simulated H-X (60 GHz)

Measured H-X

Simulated H-X

49



-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330



50


-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330



51



-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330



52


-55

-45

-35

-25

-15

-5

50

30

60

90

120

150

180

210

240

270

300

330



53

Gain

Figure 48: Gain of Design 4 (measurement and simulation)

3.4.3 Results summarized

The measurement patterns at 55 GHz, 60 GHz and 65 GHz show asymmetric E-Co patterns.

The asymmetric E-Co also appear in simulation but not as serious as in the measured results.

Radiation loss to free space from the transmission line is the prime suspect of the poor E-Co.

The following figure shows that on the E-plane, E-fields are radiated from the transmission

line so that less power is fed to the VPA in the range 0 < p < 30°.

0

2

4

6

8

10

12

14

16

55.00 60.00 65.00 70.00 75.00

Ga

in (

dB

i)

Frequency (GHz)

Simulated gain

Measured gain

54

Figure 49: Explanation for the unbalanced E-X

Besides the degraded E-Co, other performances of the antenna are satisfactory. A wide

bandwidth of 15 GHz is attained with central frequency at about 59 GHz. Antenna gain above

8 dBi is achieved from 60 GHz to 73.5 GHz. The antenna therefore meets the 802.11 ad

requirements.

As a whole, the above study shows that a 60 GHz antenna can be fabricated using VPA. The

remaining problem lies in the feeding method. It is important to devise some effective way to

feed the antenna with minimal radiation loss to free space.

In the next section, the antenna operation mechanism will be explained, some of the measurement results will be further investigated in detail and application of the antenna will be discussed.

55

4 Chapter Four Discussions

As this antenna is in VPA structure, it is worthwhile to study the operation mechanism of the

VPA.

4.1 An attempt to develop a theoretical VPA model

VPA is developed from patch antenna. In a patch antenna, it is the fringing fields that are

responsible for radiation.

Figure 50: Radiation of a patch antenna

The fringing effect varies with the dielectric constant of the substrate /0. The smaller the

value of /0, the greater is the fringing effect and hence higher radiation.

A VPA can be considered as a stack of annular patch antenna with infinitesimal difference

between inner and outer radius (q → 0 as shown in (a) of the following diagram). Free

charges reside on the wall of the VPA. These charges interact with the charges on the ground

and fringing E-fields exist between the VPA and the ground. Regarding VPA as a continuous

summation (integration) of annular patch antennas with infinitesimal thickness (differential

thickness) and each layer of patch antenna has its radiation, the total radiation of the whole

structure is the integration of radiation of all layers of patch antenna (as shown in (b) of the

56

following diagram). Therefore the radiation intensity of a VPA is greater than that of a patch

antenna and the total E-field is enhanced, resulting in higher gain of a VPA.

Figure 51: Annular patch antenna developed to circular VPA

It is intended here to find a model for the central frequency of a circular VPA. Given that the

VPA is evolved from annular patch antenna, methods in analyzing annular patch antenna are

used as a reference. According to Schneider (1969), the cutoff frequency of an annular patch

antenna is

kr6 = sr6$2Ftuv:

Where ( – angular mode number o – radial mode number $ – velocity of light

57

v: = 12 (v0 + 1) + 12 (v0 − 1)w1 + 10xyz + 34 x{ − yt − 34 x{|

}��

x – thickness of the dielectric t – inner radius z – outer radius sr6 – Eigenvalues of the modes or zeros of Bessel function (the eigenvalues and the zeros

are list below for reference

1 2.4048 3.8317 5.1356 6.3802 7.5883 8.7715

2 5.5201 7.0156 8.4172 9.7610 11.0647 12.3386

3 8.6537 10.1735 11.6198 13.0152 14.3725 15.7002

4 11.7915 13.3237 14.7960 16.2235 17.6160 18.9801

5 14.9309 16.4706 17.9598 19.4094 20.8269 22.2178

Table 5: Zeros for Bessel function of first kind

( sr� sr� sr-

0 2.405 5.520 8.654

1 3.832 7.016 10.174

2 5.135 8.417 11.620

Table 6: Eigenvalues for circular modes

Applying the above formula to a VPA, the resonance frequency will be revised to

kr6 = sr6$2F~uv:

Where ~ – physical radius of the VPA

58

v: = 12 (v0 + 1) + 12 (v0 − 1) R1 + 203 S}��

In this project, ~ = 0.65mm and v: = 1.81.

The air cavity radius of the fabricated antenna is 0.65 mm. Applying the above equation, the

resonance frequency in the fundamental mode is 131 GHz, which is not agreeable with the

lowest measured resonances of 52.5 GHz. This discrepancy may be due to the stronger

fringing E-fields induced in VPA than those in an annular patch antenna. With stronger

fringing E-fields, the effective radius ~:;; is a larger than the physical radius. Suppose the

measured resonance of 52.5 GHz is the fundamental mode,~:;; would be estimated to be

1.485 mm. As resonances are also observed at 55 GHz and 64 GHz, the corresponding ~:;;

are listed as follows:

Frequency (GHz) Effective VPA radius ~:;; (mm) Mode 52.5 1.626 ��#� 55 2.473 �� 64 2.847 �� Table 7: Effective radius at different frequencies

Stronger fringing E-fields induced in a VPA than an annular patch antenna result in the

higher radiation intensity and hence the gain of a VPA. The above effort finds that the central

frequency varies with the effective radius, but the effective radius is a hypothetical value

instead of a physical dimension. The study here fails to incorporate the physical height of the

VPA in the model. The VPA height is wrongly treated as the dielectric thickness in the above

calculation instead of the parameter. This model evolved from Schneider’s one cannot be

completed here.

59

4.2 Explaining the discrepancies between measured and simulated resonances and bandwidths

The measured reflection coefficients show three resonances as different from one resonance

in the simulation results. The measured bandwidth is also broadened.

Such discrepancy is due to the difference between experimental set up and simulation. In

experimental set up, there is air gap between the connector and the fixture on which the

antenna is rest. There is also air gap between the substrates. However such air gaps are not

accounted for in the simulation. Dielectric constant of the air is 1, which is lower than that of

the Duroid. This means E-fields are more penetrable to air than Duroid. Therefore with the

air gaps, more power is transmitted from the coaxial cable to the antenna. The air gaps

provide additional resistive load and absorb energy. This lowers the reflection coefficient.

The whole reflection coefficient curve is shifted down and increases the bandwidth.

To eliminate the effect of the air gaps, the standard of –12 dB can be used for bandwidth

measurement. By considering the –12 dB bandwidth, the bandwidth for the measured

reflection coefficient is approximately 8 GHz. Its shape follows that of the simulated

reflection coefficient and the resonances at 52.5 GHz and 55 GHz are out of the band.

4.3 Satisfying the 802.11 ad central frequency and bandwidth requirements

The central frequency of the VPA is 59 GHz which is close to the WiGig central frequency of

60 GHz.

60

Figure 52: 60 GHz band spectrum in different countries4

The 60 GHz bands available for WiGig communication in different countries are shown in

the above figure. The bands extend from 57 GHz to 66 GHz while the antenna developed in

this project covers the range from 52 GHz to 66 GHz. Thus the antenna can be adopted in

these countries as a communication device.

4 Wireless Gigabit Alliance, “WiGig white paper defining the future multi-gigabit wireless communications”,

July 2010.

61

4.4 Radiation patterns explained

The following radiation patterns are stated as follows to facilitate the discussions.

Figure 53: Radiation patterns defined

62

4.4.1 E-plane Co-polar

To explain the radiation patterns, ��#� is used as an example. In ��#� mode, the E-field strength within the air substrate enclosed by the VPA follows the Bessel function �#. Based on the graph of �#, it is deduced that the field strength is the strongest at the center and diminishes to zero on the wall of the VPA. The field strength is originated from the fringing E-fields caused by the surface charges on the wall of the VPA.

Figure 54: Bessel function of first kind

The fields are visualized in the following figure. The E-Co pattern is the gain along the p

direction on the y-z plane. If there is a receiving antenna placed on the y-z plane and oriented

along the p axis, the incident E-fields on the receiving antenna will be maximal at p = 0° and minimal at p = 90°. This is the E-Co.

63

Figure 55: E-plane Co-polarization

64

4.4.2 H-plane Co-polar

Current arriving the VPA from the feed line will flow along the circumference of the VPA as

surface current. According to Ampere’s law, there are H-fields enclosing the current. Also the

changing H-fields incur E-fields whose polarization is orthogonal to H-fields. It is therefore

deduced that the total E-fields consist of �-component. As H-fields are the strongest on the

H-plane (x-z plane), ��will have maximum strength on the H-plane. The current flow, H-

fields and E-fields are demonstrated in the following diagram.

Figure 56: H-plane Co-polarization

65

The H-Co pattern is the gain along the � direction on the x-z plane. If there is a receiving

antenna placed on the x-z plane and oriented along the � axis, its received power at different

elevation angles p is H-Co.

4.4.1 E-plane Cross-polar (explaining the large measured E-X)

The E-X pattern is the gain along the � direction on the y-z plane. Ideally the gain should be

zero. However currents flow continuously along the VPA circumference, even at a point on

the y-z plane. Therefore H-fields exist on E-plane and hence the E-fields which are directed

along the � axis. This explains the large E-X as observed in the measured patterns.

Figure 57: E-plane Cross-polarization

66

4.4.1 H-plane Cross-polar (explaining the large measured H-X)

The H-X pattern is the gain along the p direction on the x-z plane. The H-X should be zero in

an ideal case. However there are surface charges on the wall of the antenna, fringing fields

are directed to or from the ground to every point on the VPA wall. It is therefore the p-

component of the E-field or ��exists on every vertical plane containing the z-axis, including

the H-plane. This results in large H-X patterns as observed on the measured patterns.

However currents flow continuously along the VPA circumference.

Figure 58: H-plane Cross-polarization

In fact the large E-X and H-X are the drawbacks of VPA.

67

4.5 Large null at �~��° on E-Co

The unbalanced shape of E-Co has been explained extensively in previous sections. It is

mainly caused by asymmetric antenna structure posed by the transmission line. However

there is one more problem with the E-Co. That is the large null at p~315°. Explanation can

be sought in the paper by Wong et al (2010) based on which the antenna of this project is

developed Their VPA at 33 GHz also exhibits a serious null at p~315° as shown in the

following figure.

Figure 59: E-plane polarization of the VPA at 33 GHz5

According to the paper, the null is due to the radiation from the probe of the SMA connector.

The following diagram shows how the antenna is connected to the SMA connector. Before

measurements can be taken, a conducting pin is soldered on the transmission line. The pin is

then inserted to the probe of an SMA connector which will be in turn connected to a network

analyzer by an RF cable. As the pin and the probe are conducting materials, E-fields are

radiated from them. The radiation may be directed towards the angle θ~315° and cancel part

of the radiation from the VPA. As the VPA of this project is measured in the same way, the

radiating fields from the conducting pin are expected to be the prime cause for the null.

5 H. Wong, K. B. Ng, K. M. Luk, C. H. Chan and Q. Xue, “Printed millimeter wave vertical patch antenna,” IEEE Conference publications,

pp. 647-649, 2010.

68

Figure 60: Radiation caused by an SMA connector

4.6 Orientation of the antenna in application

According to the radiation patterns, the antenna should be wall-mounted.

4.6.1 E-plane: received power and vertical beamwidth

Suppose the wall is parallel with the x-y plane and the vertically polarized incoming signal

aligns with the y-axis. The following diagram shows how the incoming signal on the E-plane

is received by the antenna. ��(p) and ��(p) are incoming signals with incident angles p = 0° and p > 0° respectively. Both are vertically polarized. The measured E-Co and E-X at

60 GHz are also shown for reference. At p = 0°, �� is parallel with the incoming signal, so

the signal ��(p), in theory, can be well received by the antenna. Measurement results show

that the E-Co at p = 0° is −2.625XH, that is more than half-power is received. However the

vertical beamwidth is hardly more than �10°. ��(p) may not be strong enough to induce a

69

voltage on the antenna. The antenna may not be able to receive incoming signals with

inclination angle more than �10°.

Figure 61: Received signal on E-plane

4.6.2 H-plane: received power and horizontal beamwidth

The H-plane is then considered. In the following figure, the antenna is rotated and viewed

from a different perspective. ��-(p) and ��(p) are incoming signals with incident angles p = 0° and 30 > p > 0° respectively. Similar to the analysis for the E-plane, �� atp = 0° is

parallel with the incoming signal, so the signal ��(p), in theory, can be well received by the

antenna. Measurement results show that the H-Co at p = 0° is −0.156XH, that is more than

70

half-power is received. The horizontal beamwidth is about than �30°. So as long as ��(p) incidents on the antenna in less than �30°, it can be detected by the antenna.

Figure 62: Received signal on H-plane

4.7 Beamwidth for WLAN applications

802.11 ad is expected to be widely used for WLAN. For WLAN applications, the beamwidth

should be as wide as possible, and ideally the antenna should be omnidirectional. It is

because a device in WLAN setting is expected to receive signals from the surrounding at any

incident angle. If an antenna with a narrow beamwidth is used, some signals may not be

detected. In this regard, the VPA with horizontal beamwidth of �30° may not be a preferred

choice for WLAN applications. However this antenna can be used for point-to-point wireless

connection.

71

4.8 Difference between co-polarization and cross-polarization

At central frequency, the X-polar gain is 19 dB below the Co-polar on H-plane and the

difference is 18 dB for E-plane at p = 0°. The radiation patterns show that such difference is

larger than 20 dB for the range �30° for H-plane. Referring to the VPA by Wong et al (2010),

the Co-polar and X-polar difference is around 20 dB. Adopting this as benchmark, the

performance of the VPA developed in this project is comparable to that by Wong et al (2010).

4.9 Antenna performances at other frequencies

Performances of the antenna at different frequencies are listed below.

Normalized gain (dB) p = 330° p = 0° p = 30° E-Co −1.145 −7.201 −2.821 E-X −24.174 −14.273 −15.429 H-Co −1.253 −5.022 −2.523 H-X −18.322 −25.752 −21.057 Vertical beamwidth: - Horizontal beamwidth: More than �30° |E-Co| – |E-X|: 7.072 dB at p = 0° |H-Co| – |H-X|: 20.73 dB at p = 0° Table 8: Antenna performance at 55 GHz

Normalized gain (dB) p = 330° p = 0° p = 30° E-Co −1.801 −2.625 −14.625 E-X −36.964 −21.412 −35.527 H-Co −2.453 −0.156 −1.568 H-X −29.672 −19.745 −27.527 Vertical beamwidth: Less than �10° Horizontal beamwidth: �30° |E-Co| – |E-X|: 18.787 dB at p = 0° |H-Co| – |H-X|: 19.589 dB at p = 0° Table 9: Antenna performance at 60 GHz

72

Normalized gain (dB) p = 330° p = 0° p = 30° E-Co −8.676 −7.46 −11.666 E-X −31.733 −32.626 −20.45 H-Co −1.922 −0.751 −1.603 H-X −26.356 −16.429 −21.211 Vertical beamwidth: - Horizontal beamwidth: More than �30° |E-Co| – |E-X|: 25.166 dB at p = 0° |H-Co| – |H-X|: 15.678 dB at p = 0° Table 10: Antenna performance at 65 GHz

Overall speaking, horizontal beamwidth is at least �30° at all frequencies and the lowest Co-

polar and X-polar differences are 7.072 dB for E-plane and 15.678 dB for H-plane

respectively.

73

4.10 Gain

Measurement results show a gain of at least 8 dBi for the range from 60 GHz to 73.5 GHz

and the average gain is 8.65 dBi. The following table shows the gains of other antennas. The

antennas are ranked in descending order according to their gain. The gains range from –19

dBi to 25 dBi. The VPA developed in this project is in the middle of the range. Its gain of

8.65 dBi is therefore acceptable.

Literature Antenna type Gain (dBi) GHz Pavuluri et al (2008) FSS based cavity 25 60 Karnfelt et al (2006) Microstrip array 18 60

Sironen, Qian and Itoh (2001) Conical horn antenna 16.5 60

Zhang and Sun (2009) Grid array antenna 15 60 Zhang, Sun and Guo (2005). Quasi-Yagi 12.5 60

Grzyb et al (2006) Suspended antenna 8.5 60 Mak, Luk and Lee (2003) VPA 8 5.6

Wong et al (2010) VPA 7.9 33

Liu et al (2009) Aperture-coupled patch antenna 7 60 Lau et al (2006) VPA 7 7

Lau et al (2006) VPA 7 4.25 Sun et al. (2008) Yagi antenna 6 60

Lin et al (2007) triangular monopole antenna 0 60 Gutierrez et al (2009) Half-wave dipole -0.2 60

Gutierrez et al (2009) Rhombic antennas -3.55 60

Gutierrez et al (2009) Yagi -7.5 60 Zhang, Sun and Guo (2005). Yagi -12.5 60

Zhang, Sun and Guo (2005). inverted-F antenna -19 60 Table 11: Gains of other antennas

74

4.11 Air cavity (explaining how gain is amplified)

Surface loss is a problem with patch antenna which is the loss of input power when the E-

fields pass from the conducting material to dielectric. The lowered input power reduces the

gain. The air cavity embedded in the lower substrate is thus for enhancing the gain.

To attain gain enhancement, standing waves need to be established within the air cavity. The

condition for standing waves in an air cavity is 2n = oλ, where D is the diameter of the air

cavity and o is any nonzero integer. This condition is so set because the E-fields have to be

return to its initial position after travelling to and fro the air cavity. The total distance that the

E-fields travel before returning to the initial position is 2D. When the E-fields incident on the

wall of the air cavity, it is totally reflected as the wall is a conducting surface which can be

treated as short circuit, the reflection coefficient of which is –1. The total negative reflection

forms the standing waves for E-fields. With total reflection, the E-fields travel back and forth

within the air cavity, forming a virtual infinite ground plane. The virtual ground plane reflects

the radiation from the VPA and thus gain is enhanced.

Figure 63: Standing waves

The E-fields radiation from the air cavity is also shown.

75

Figure 64: Reflected E-fields from the air cavity

As a convention, the air cavity diameter is about 1 to 1.5 times of the wavelength. In this

project, the air cavity diameter is 4.74 mm and λ is 5mm. This shows that 2n =9.48((~2λ = 10mm.

76

5 Chapter Five Further research

It is stated at the end of Chapter Three that the transmission line is mainly accountable for the

asymmetric E-Co pattern. Hence further effort is put to design another feed line in order to

improve the antenna performance.

5.1 Feeding design 1 (CPW)

In this new design, a CPW feed is adopted to replace the microstrip line. With an additional

ground placed on the top of the upper substrate, E-fields are directed to the top ground

instead of radiating to free space.

Figure 65: E-field directed to the top ground

77

Figure 66: Top view of Feeding design 1

78

Figure 67: Reflection coefficient of Feeding design 1

Simulation results on reflection coefficient show that the bandwidth is 5 GHz and ripples

appear across the band. The ripples may be due to the reflections between the VPA and the

additional top ground as shown in the following figure.

Figure 68: Explanation for the ripples

79

5.2 Feeding design 2 (CPW with smaller top ground)

To reduce the reflection between the VPA and the top ground, the area of the top ground is

reduced.

Figure 69: Top view of Feeding design 2

80


81

Figure 71: Radiation patterns of Feeding design 2

As noted from the simulation results, there is no more ripple. However there is a sudden

impedance mismatch at 62.5 GHz. Refer to the radiation pattern, it is found that the problem

of asymmetric E-Co does not improve. Loss occurs at about 20°.

5.3 Feeding design 3 (Probe feed)

Probe feed is known for its wide bandwidth and symmetric patterns at lower band, so probe

feed is adopted here.

82

Figure 72: Feeding design 3 – probe feed


83


According to the simulated reflection coefficient, a wide bandwidth is obtained as expected.

However the radiation patterns are worse than previous designs. One possible reason for

symmetric radiation patterns not able to be obtained with probe feed at 60 GHz is that the

probe size is comparable to the VPA ring size. The VPA radius is 0.65 mm and the probe

radius is 0.15 mm. That is the probe radius is more than one-tenth of the VPA radius. At

lower frequency where the ring radius is the multiples of the probe radius, the radiation from

the probe is practically negligible compared to that from the VPA, so the pattern is not much

affected. At high frequency, radiation from the probe can significantly affect the E-fields

radiated from the VPA. While E-fields from both radiating elements interact with each other,

some of the fields may be cancelled, resulting in distortion and lower gain.

84

5.4 Feeding design 4 (Probe feed with transmission line)

In this design, the feed line is implemented by a transmission and a probe. The purpose is to

put the probe away from the VPA such that the radiation from the probe does not affect those

from the VPA.

Figure 75: Feeding design 4 – probe feed with transmission line

85


86


In this design, bandwidth is narrowed and ripples appear. The ripples may be due to the

reflection between the transmission line and the probe. It is also tried to move the probe

towards the VPA but similar patterns are resulted.

87

5.5 A framework for further research

After a few trials, it is still not successful to attain a symmetric pattern. However some

important points can be drawn for further research. The designs proposed above are based on

the rationale that radiation from the transmission line should be suppressed. Actually there is

already existing research in the areas of vertical transition such as feeding the antenna by a

number of vias in a multilayer structure, aperture-coupled transition and cavity-coupled

transition by Huang and Wu (2012), and CPW-to-CPW vertical transition by Enayati,

Vandenbosh and De Raedt (2010). However all these designs are rather complex in structure

while it is desirable to keep the simple structure of the VPA. Therefore the challenge ahead is

to design a planar feed line in which radiation should be directed to the ground instead of the

free space.

88

6 Chapter Six Conclusions

As a novel attempt to develop a VPA at 60 GHz, this project ends with partial success. The

antenna developed has a center frequency at 59 GHz and its bandwidth ranging from 51.5

GHz to 66.5 GHz covers the WiGig bands in all countries in which WiGig 802.11 ad is

adopted for applications. An average gain of 8.65 dBi is also achieved. The major drawback

of this antenna lies in the E-Co patterns which is asymmetric. The measured E-Co patterns

show that larger losses occur on one side of the antenna than the other. A number of nulls

also appear especially at p = 315°, which means the transmitted power is more susceptible to

losses at certain angles. The radiation loss induced in the microstrip transmission line is

suspected to be the prime cause for the asymmetric E-Co. Therefore designing an effective

feeding method is essentially inevitable for achieving a desirable radiation pattern. As a

framework for further research, the revised feed line should be less prone to free space

radiation while keeping the whole antenna structure simple. The simplicity in structure

ensures an affordable production cost for the antenna to be widely accepted in the markets.

89

References [1] CTimes http://www.ctimes.com.tw

[2] S. A. Long, L. C. Shen, D. H Schaubert and F. G. Farrar, “An experimental study of the circular-polarized elliptical printed-circuit antenna,” IEEE Transactions and antenna and propagation, vol. AP-29, No 1, pp.95-99, January 1981.

[3] Agilent Technologies, “Wireless LAN at 60 GHz – IEEE 802.11ad Explained”.

[4] D. Liu, J. Akkermans and B. Floyd, “A superstrate patch antenna for 60GHz applications”, 3rd European Conference on Antennas and Propagation EuCAP, March 2009.

[5] Y. P. Zhang and M. Sun, “An overview of recent array design for highly-integrated 60-GHz radios”, 3rd European Conference on Antennas and Propagation 2009, pp.3783-3786, IEEE 2009.

[6] M. Sun, Y. P. Zhang, K. M. Chua, L. L. Wai, D. Li, and B. P. Gaucher, “Integrated of Yagi antenna in LTCC package for differential 60-GHz radio”, ,” IEEE Transactions and antenna and propagation, vol. 56, No 8, pp.2780-2783, August 2008.

[7] J. Grzyb, U. Pfeiffer and B. Gaucher, “Wideband cavity-backed folded dipole superstrate antenna for 60 GHz applications”, 2006 IEEE and Propagation Society International Symposium 2006, pp. 3939-3942, 2006.

[8] J. Kim, H. S. Lee, H Lee and J. Yoon, “60-GHz CPW-fed post-supported patch antenna using micromachining technology”, IEEE Microwave and wireless components letters, vol.15, No. 10, pp.635-637, October 2005.

[9] J. Park and Y. Wang, “A 60 GHz integrated antenna array for high-speed digital beamforming applications”, 2003 IEEE MTT-S International Microwave Symposium Digest 2003, vol.3, pp.1677-1680, 2003.

[10] C. C. Lin, S. S. Hsu, C, Y. Hsu, and H. R. Chuang, “A 60-GHz millimeter wave CMO RFIC-on-Chip triangular monopole antenna for WPAN applications”, 2007 IEEE and Propagation Society International Symposium, pp. 2522-2525.

[11] Y. P. Zhang, M. Sun, and L. H. Guo, “On-chip antennas for 60-GHz radios in silicon technology”, IEEE Transactions on electron devices, vol. 52, No. 7, pp.1664-1668, July 2005.

[12] F. Gutierrez, Jr., K. Paarrish, and S. Rappaport, “On-chip integrated antenna structure in CMOS for 60 GHz WPAN systems”, IEEE Journal on selected areas in communications, vol. 27, No. 8, pp1367-1378, October 2009.

[13] M. Sironen, Y. Qian and T. Itoh, “A 60GHz Conical horn antenna excited with Quasi-Yagi antenna,” 2001 IEEE MTT-S International Microwave Symposium Digest, vol.1, pp.547-550, 2001.

90

[14] C. Karnfelt, P. Hallbjorner, H. Zirath and A. Aliping, “High gain active microstrip antenna for 60GHz WLAN/WPAN application,” IEEE Transactions on Microwave Theory and Technology, vol. 54, No. 6, June 2006.

[15] S. K. Pavuluri, A.P. Feresidis, G. Gouseetis, and C.H. Wang, “A novel high gain 60GHz antenna using a suspended frequency selective surface (FSS)”, Metamaterials, September 2008.

[16] C.L. Mak, K.M. Luk, and K.F. Lee, “Geometry of wideband small-size antenna: Vertical patch antenna,” Elec. Lett., vol. 39, pp.1777-1779, 2003

[17] K.M. Lau, H. Wong, C.L. Mak, K.M. Luk, and K.F. Lee, “A vertical patch antenna for dual-band operation,” IEEE Antenna and wireless propagation letters, vol. 5, pp.95-97, 2006

[18] H. Wong, K. B. Ng, K. M. Luk, C. H. Chan and Q. Xue, “Printed millimeter wave vertical patch antenna,” 2010 International conference on communications, circuits and systems (ICCCAS), pp. 647-649, 2010.

[19] M.V. Schneider, “Microstrip lines for microwave integrated circuits”, Bell System Technical Journal, 48, pp.1421-1444, 1969.

[20] Wireless Gigabit Alliance, “WiGig white paper defining the future multi-gigabit wireless communications”, July 2010.

[21] X. Huang and K. L. Wu, “A broadband and vialess vertical microstrip-to-microstrip transition”, IEEE Transactions on microwave theory and techniques, vol. 60, No. 4, pp.938-944, April 2012.

[22] A. Enayati, G. A. E. Vandenbosch, and W. De Raedt, “CPW-to CPW transitions for 3D packaging and antenna measurement at millimeter wave frequencies”, 32nd ESA antenna workshop, 5-8 October 2010.

[23] S. Sendra, M. Garcia, C. Turro, and J. Lloret, “WLAN IEEE 802.11a/b/g/n indoor coverage and interference performance study”, International journal on advances in networks and services, vol. 4, No 1&2, pp.209-222, 2011.

[24] Y. P. Zhang, M. Sun, K. M. Chua, L. L. Wai, and D. Liu, “Antenna-in-package design for wirebond interconnection to highly integrated 60-GHz radios”, IEEE Transactions and antenna and propagation, vol. 57, No 10, pp.2482-2851, October 2009.

this document is downloaded from cityu …lbms03.cityu.edu.hk/oaps/ee2013-fyp-lsy001.pdfthis...

Documents