antenna designing

5/24/2018 Antenna Designing

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Advances in Low-Profile Antennas

in Wireless Communications

Guest Editors: Guo Qing Luo, Xiao Ping Chen,

Zhang Cheng Hao, Bing Liu, and Yu Jian Cheng

International Journal of Antennas and Propagation


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Advances in Low-Profile Antennas in

Wireless Communications


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Advances in Low-Profile Antennas in

Wireless Communications

Guest Editors: Guo Qing Luo, Xiao Ping Chen,Zhang Cheng Hao, Bing Liu, and Yu Jian Cheng


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Copyright Hindawi Publishing Corporation. All rights reserved.

Tis is a special issue published in International Journal of Antennas and Propagation. All articles are open access articles distributedunder the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, pro-

vided the original work is properly cited.


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Editorial Board

Mohammod Ali, USACharles Bunting, USAFelipe Catedra, SpainDau-Chyrh Chang, aiwanDeb Chatterjee, USAZ. N. Chen, SingaporeMichael Yan Wah Chia, SingaporeShyh-Jong Chung, aiwanLorenzo Crocco, Italyayeb A. Denidni, CanadaKaru Esselle, AustraliaFrancisco Falcone, SpainMiguel Ferrando, SpainVincenzo Galdi, ItalyWei Hong, Chinaamer S. Ibrahim, USANemai Karmakar, AustraliaSe-Yun Kim, Republic of Korea

Ahmed A. Kishk, CanadaSelvan . Krishnasamy, IndiaJu-Hong Lee, aiwanByungje Lee, Republic of KoreaJoshua Le-Wei Li, ChinaYilong Lu, SingaporeJ.S. Mandeep, MalaysiaAtsushi Mase, JapanGiuseppe Mazzarella, ItalyDerek McNamara, CanadaC. F. Mecklenbrauker, AustriaMichele Midrio, ItalyMark Mirotznik, USAAnanda S. Mohan, AustraliaP. Mohanan, IndiaPavel Nikitin, USAA. D. Panagopoulos, GreeceMatteo Pastorino, Italy

Massimiliano Pieraccini, ItalySadasiva M. Rao, USASembiam R. Rengarajan, USAAhmad Safaai-Jazi, USASaeddin Safavi-Naeini, CanadaMagdalena Salazar-Palma, SpainStefano Selleri, ItalyZhongxiang Shen, SingaporeJohn J. Shynk, USASeong-Youp Suh, USAParveen Wahid, USAYuanxun Ethan Wang, USADaniel S. Weile, USAat Soon Yeo, SingaporeYoung Joong Yoon, Republic of KoreaJong-Won Yu, Republic of KoreaWenhua Yu, USAAnping Zhao, China


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Contents

Advances in Low-Prole Antennas in Wireless Communications, Guo Qing Luo, Xiao Ping Chen,Zhang Cheng Hao, Bing Liu, and Yu Jian ChengVolume , Article ID , pages

A . GHz Cross Rhombic Antenna for a Cube Satellite Application, Jorge Sosa-Pedroza,Sergio Pena Ruiz, and Fabiola Martnez-ZunigaVolume , Article ID , pages

Novel Dual Band Microstrip Circular Patch Antennas Loaded with ENG and MNG Metamaterials,Md. Ababil Hossain, Md. Saimoom Ferdous, Shah Mahmud Hasan Chowdhury, and Md. Abdul Matin

Volume , Article ID , pages

Multishorting Pins PIFA Design for Multiband Communications, Muhammad Sajjad Ahmad, C. Y. Kim,and J. G. ParkVolume , Article ID , pages

Multiband Printed Asymmetric Dipole Antenna for LTE/WLAN Applications , Chia-Mei Peng,I-Fong Chen, and Chin-Hao LiuVolume , Article ID , pages

Microstrip Folded Dipole Antenna for GHz MMW Communication , Guang Hua, Chen Yang, Ping Lu,Hou-Xing Zhou, and Wei HongVolume , Article ID , pages

BCB-Si Based Wide Band Millimeter Wave Antenna Fed by Substrate Integrated Waveguide,Hamsakutty Vettikalladi and Majeed A. S. AlkanhalVolume , Article ID , pages

A Wireless and Real-Time Monitoring System Design for Car Networking Applications, Li Wenjun,Zhong Yiming, and Li WenbinVolume , Article ID , pages

A Compact CPW-Fed UWB Antenna with Dual Band-Notched Characteristics,Aiting Wu and Boran GuanVolume , Article ID , pages

Implementation of Low-Cost UHF RFID Reader Front-Ends with Carrier Leakage Suppression Circuit,Bin You, Bo Yang, Xuan Wen, and Liangyu QuVolume , Article ID , pages

Cavity-Backed Dipole Antenna for Intelligent Lock Communication, Bo Yuan, Wen Wang,Xiao Hong Zhang, and Guo Qing LuoVolume , Article ID , pages

Millimeter Wave on Chip Antenna Using Dogbone Shape Articial Magnetic Conductor, Guo Qing Luo,Zheng Zheng Song, Xiao Hong Zhang, and Xiao Ping HuVolume , Article ID , pages


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An AMC Backed Folded Dipole Slot Antenna Based on CMOS Process, Guo Qing Luo, Lin Qi Wu,and Xiao Hong ZhangVolume , Article ID , pages

Review of Low Prole Substrate Integrated Waveguide Cavity Backed Antennas, Guo Qing Luo,ian Yang Wang, and Xiao Hong ZhangVolume , Article ID , pages

Uniplanar Millimeter-Wave Log-Periodic Dipole Array Antenna Fed by Coplanar Waveguide,Guohua Zhai, Yong Cheng, Qiuyan Yin, Shouzheng Zhu, and Jianjun Gao

Volume , Article ID , pages

Design and Implementation of Double-Transmitter-Coil Antenna Used for the Tag Test System,Bin You, Haoling Yue, Xuan Wen, and Liangyu QuVolume , Article ID , pages

Simplied Printed Log-Periodic Dipole Array Antenna Fed by CBCPW , Guohua Zhai, Yong Cheng,Qiuyan Yin, Shouzheng Zhu, and Jianjun GaoVolume , Article ID , pages

Broadband Multilayered Array Antenna with EBG Reector, P. Chen, X. D. Yang, C. Y. Chen,and Z. H. MaVolume , Article ID , pages

A Novel Dual-Shorting Point PIFA (GSM to IMT-A) for Mobile Handsets, Pengcheng Li, Jin Pan,Deqiang Yang, and Pingzai NieVolume , Article ID , pages

A UWB Band-Pass Antenna with Triple-Notched Band Using Common Direction Rectangular

Complementary Split-Ring Resonators, Bo Yan, Di Jiang, Ruimin Xu, and Yuehang XuVolume , Article ID , pages

Substrate Integrated Waveguide Fed Cavity Backed Slot Antenna for Circularly Polarized Application ,Xiao Hong Zhang, Guo Qing Luo, and Lin Xi DongVolume , Article ID , pages


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EditorialAdvances in Low-Profile Antennas in Wireless Communications

Guo Qing Luo,1 Xiao Ping Chen,2 Zhang Cheng Hao,3 Bing Liu,4 and Yu Jian Cheng5

College of Electronic Information, Hangzhou Dianzi University, Hangzhou , China Department of Electrical Engineering, Poly-Grames Research Centre, University of Montreal, Montreal, QC, Canada HT J School of Information Science and Engineering, Southeast University, Nanjing , China College of Electronic and Information Engineering, Nanjing University of Aeronautics and Astronautics, Nanjing , China School of Electronic Engineering, University of Electronic Science and Technology of China, Chengdu , China

Correspondence should be addressed to Guo Qing Luo; [email protected]

Received January ; Accepted January ; Published February

Copyright Guo Qing Luo et al. Tis is an open access article distributed under the Creative Commons Attribution License,which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Antenna is a type of device that is adapted to transmit or

receive electromagnetic energy. Numerous differing typesof antenna structures have been developed, such as reec-tor antenna, helical antenna, notch antenna, cavity backedantenna, patch antenna, and line antenna. With rapid devel-opment of wireless communications, low-prole antenna isin great demand, especially in handheld radios, wirelessUSB, wireless sensors networks, high speed WPAN, andmobile devices. For the attractive characteristics such aslight weight, good conformability, easy integration, and lowcost fabrication, low-prole antennas have been extensivelystudied by researchers both in academia and industry. ISIindexed papers whose titles include the words low-proleand antenna in the past decades have been counted. Fromthe results shown in Figureit can be found that the researchtopic on low-prole antenna has become more and morepopular.

One low-prole antenna is the patch antenna, which isusually elevated above a large metal plate. Dielectric substrateis used to support the elevated portion of the antenna abovethe large metal ground plate. Te most popular type of patchantenna is microstrip patch antenna, which is manufacturedby printed circuit board materials and process. Te spacingbetween the top patch and the bottom metal ground planeis on the order of / wavelength and a low physical depthprole can be easily achieved. Main problems of patchantenna include a relatively narrow bandwidth, a high radi-ation angle above the horizon, and various manufacturing

and fabrication difficulties. Its bandwidth can be increased

by adding radiating surfaces and increasing the volume ofthe antenna, adding an impedance compensating network,placing selected impedance into the radiating surface, andintroducing resistances into the radiating surface, thus low-ering the Q of the antenna. Its radiation angle can be loweredby using a dielectric structure.

As a low-prole antenna, dipole antenna is also widelyused in wireless communication for its vertically polarizedradiation performance. Te arm length of a dipole is abouta half wavelength, which is too long for applications whenit operates at a low frequency. Small size antennas such asplanar inverted F antenna (PIFA), loop antenna, and tableantenna have been presented. Te PIFA is a variation of thepatch antenna, whose oneend of a radiating element disposedon a groundplane isbent so asto be connected to the groundplate. Its edge feeding structure is not easy to ush mounting.It also needs a grounded tuning wire separated from themetal sheet of the radiator. It also suffers from limitationsof a narrow bandwidth and a high elevation radiation. Asimple loop antenna is constructed by protruding the innerconductor of a coaxial line from one point of the groundplane and its end connects with the ground plane at the otherpoint. A low-prole loop antenna is hard to match for itshigh capacitive impedance and low radiation resistance. Atable antenna constructed by a radiating plate supported byfour conductor posts was disposed on the ground plane. Acoaxial feed is connected to the central part of the radiating


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element and a broad bandwidth can be achieved. If its heightis reduced to get low-prole, the size of its radiating platemust be increased.

In modern wireless communication system, the designconsiderations of antenna include not only low-prole, highradiation performance, but also compactness, low cost, andeasy integration. Recently antennas integrated on board,in package, and on chip have been extensively studied.Tis special issue presents novel antenna designs, antennaminiaturization and optimization techniques, antenna per-formance improvement techniques, feeding mechanisms andantenna arrays, and antenna applications.

We hope that through this special issue, the readers

will nd not only new designs about different low-proleantennas but also their valuable applications.

Guo Qing LuoXiao Ping Chen

Zhang Cheng HaoBing Liu

Yu Jian Cheng


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Research ArticleA 2.4 GHz Cross Rhombic Antenna fora Cube Satellite Application

Jorge Sosa-Pedroza, Sergio Pea Ruiz, and Fabiola Martnez-Ziga

Unidad Profesional Adolfo Lopez Mateos, Escuela Superior de Ingeniera Mecanica y Electrica, Instituto Politecnico Nacional,Edicio Z-, er. Piso, Colonia Lindavista, Mexico, DF, Mexico

Correspondence should be addressed to Sergio Pena Ruiz; [email protected]

Received August ; Revised December ; Accepted December ; Published February

Academic Editor: Xiao Ping Chen

Copyright Jorge Sosa-Pedroza et al. Tis is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

We present designand construction results o a .Ghz cross rhombic antenna to be used in a cubesat. Computationaldesign agreeswith experimental results afer its construction. cross rhombic antenna is a novel planar structure o our own design, presentingcircular polarization and medium gain; it is built over a RFA substrate with

= 6.15, decreasing its size to t the required

dimensions o satellite. A special characteristic o this design is the enhancing o operational bandwidth using a technique we havebeen studying, related to sofening the structures with sharp edges. Results show applicability and success o our technique.

1. Introduction

A cube satellite is a very small spacecraf, usually having an

area as much as cm2 and no much longer than cm;its weight should be less than Kg. Most o these smallsatellites use commercial electronic components. CaliorniaPolytechnic State University and Stanord University issuedthe CubeSat specications in , helping universities todevelop the science and space exploration; these specica-tions are ollowed by amateur radio satellite builders. It is

easy to imagine that the cm2 satellite area requires anantenna to t those dimensions. We have been working with

planar antennas or long time, developing, designing andconstructing our own structures []. Some o them have beenproposed to be used in different applications. We presentin this paper a study o one o those structures: the CrossRhombic Antenna (CRA). Actually, there are many kindso planar antennas having different orms. Planar antennasare very popular considering their small dimensions and lowcost making them ideal or many applications where smalldimensions are needed, as in airplanes, spacecraf, and xedterrestrial communications taking a great importance in lastgeneration o these systems. Some characteristics to be con-sidered or the antenna design are the ollowing:

(i) high reliability;

(ii) small size;

(iii) low weight;

(iv) high efficiency;

(v) low cost.

Te cubesat dimensions impose as well antenna dimen-sions, keeping the best behavior. As we show in Section ,

analysiswas done in order toget the bestbehavioror a cm2

o satellite ace. CRA is proposed to be used in Sensat, acube satellite, working at . GHz. Sensat is a small educa-

tional satellite projected with optical and microwave pay-loads, developed by several Mexican universities.

2. The Cross Rhombic Antenna

CRA belongs to the amily o traveling wave antennas, withcircular polarization, medium gain, and a directional eldpattern. Antenna is ed in one end and loaded in the otherone, to meet expected behavior. Te parametric analysiswe present takes in account different characteristics o theantenna such as: microstrip width, load impedance, type osubstrate and substrate thickness as well as sofening o sharpcorners. Antenna behavior is related to its gain, eld pattern,axial ratio, and 50 input coupling; requency design was


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International Journal o Antennas and Propagation

Feed Load impedance

Truncated arm

Interior arm

Exterior arm

Maximum length

a

F : Cross rhombic antenna.

. GHz and we use antenna, Computer Simulation ech-

nology (CS) simulator. Modiying structure characteristicsusing a RFA substrate with

= 6.15, we select the

best one to construct an antenna prototype to compare itwith simulation results. CRA is an evolution o the antennaproposed by Roederer in []. Antenna is a rhomb shapemicrostrip over a ground plane as shown in Figure . Circularright or lef polarization depends on eed and load position.Load impedance could be any value but we have noticed thatthe best results are obtained using short circuit, open circuitor .

3. Parametric Analysis Using RF60A Substrate

In this section, simulation results are presented, modiyingphysical characteristics o the RFA substrate structurehaving

= 6.15 and = 0.0504 m at . GHz. At this

requency, substrate width is = /11. Original designconsiders the maximum dimension (Figure ) as 1.67 or

. cm, matching the cm2 area o cube satellite. Physicalparameters considered were as ollows: microstrip width,load impedance, substrate thickness, and sofening o sharpcorners.

.. Microstrip Line Width. First case o analysis wasthe microstrip line widths. able shows the relationshipbetween gain and line width, or the three selected loadimpedances, while Figures , , and show the relatedmicrostrip line width to return loss parameter or each load.

As seen inable , the best gain results are obtained or3

= 0.033 = 1.7 mm with open circuit load, but we choose5

= 0.041 = 2.1 mm, considering the better couplingresults oFigure ; however, as seen in Figure , there isa better coupling or

6 and

7but at higher requencies.

Figure shows a better coupling with thicker lines or the50 load, but the gain is lower, as seen inable . For shortcircuit load, gain is similar to that o open circuit load andbetter coupling but at higher requency; to reduce couplingrequency is necessary to increase antenna dimensions whichis not desirable. Considering these results, we choose . mm

2.35

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F : Return losses (open load line analysis).

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a1a2a3a4

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F : Return losses ( load line analysis).

or microstrip line width and open circuit load. Figure shows eld patterns ( = 9 0 and = 180) and we cansee similarity between all o them, and then they are not aparameter to be used or comparison.

4. Further Size Reduction

We tried a urther size reduction o RCA, moving cornersto the antenna center, but with little changes in microstriplines width, as shown in Figure . able shows the naldimensions. o dene the new microstrip width, we analyzethe effects on gain and coupling; results are shown inable andFigure .Figure depicts the different eld patterns oreach microstrip width.


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Return

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(dB)

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a1a2a3a4

a5a6a7

F : Return losses (short load line analysis).

: Line width analysis (RFA).

Width Width Gain open Gain Gain short

(mm) () (dBi) (dBi) (dBi)

1

= 1.3 . . . .

2

= 1.5 . . . .

3

= 1.7 . . . .

4

= 1.9 . . . .

5

= 2.1 . . . .

6 = 2.3 . . . .7

= 2.5 . . . .

: Antenna dimensions (RFA).

Original Modied

Maximum length . .

Interior arm . .

Exterior arm . .

runcated arm . .

Angle between interior arms () . .

Angle between exterior arms () . .

Angle interior-exterior arms () . .

5. Substrate Thickness

Although we can make any change in dimensional antennacharacteristics in simulation process, a limitation arises iwe want to construct it, because commercial substrates havedened thickness dimensions; then we change that dimen-sion in order to know its effects on radiation parameters,taking special attention in results with the materials onhand. Using the best ormer results, simulation was made ordifferent substrate thicknesses, as able , shows with gainresults or each one. Figure depicts the coupling results.

: Antenna gain line analysis.

Width (mm) Gain (dBi)

1

= 1.5 .

2

= 1.6 .

3

= 1.7 .

4 = 1.8 .5

= 1.9 .

6

= 2.0 .

7

= 2.1 .

: Antenna gain substrate thickness analysis.

Tickness (mm) Gain open (dBi)

1

(/5) = 10 .

2

(/6) = 8.4 .

3

(/7) = 7.2 .

4

(/8) = 6.3 .

5

(/9) = 5.6 .

6

(/10) = 5.04 .

7

(/11) = 4.58 .

8

(/12) = 4.2 .

9

(/13) = 3.8 .

10

(/14) = 3.6 .

Figure depicts the coupling results, as seen = /11 hasagain the best coupling results, but the commercial thicknesso substrate we have is = /9 = 5.6 mm. We keep theanalysis o both the thickness substrates, in the ollowingparagraphs, as a way o comparison.

6. Softening Sharp Edges toEnhance Bandwidth

It can be noticed in 11

parameters o ormer gures thatcurves are very sharp with small bandwidth, making antennadesign and construction with a high dependence on re-quency; repeatability could be a problem considering highdependency o antenna parameters o structure dimensions.o avoid this problem, we propose to increase bandwidthsofening antenna corners, making it less dependent onrequency [, ]; to set those changes we selected the bestparameters o the analysis, that is, strip line width =2.0 mm, substrate thickness = /11, and open circuit load;antenna modication is shown inFigure .

We noticed a small change in requency coupling afersofening edges; to solve this problem, all dimensions weremodied and a new substrate thickness analysis was done inorder to get a new optimum value or at the . GHz res-onant requency; simulation results are presented inable andFigure . Good coupling or

4 = /8 and

5 = /9is

noticed, and the last one is precisely the thickness we haveto construct; Figures andshow eld patterns or thosethicknesses.

Final optimized dimensions are presented inable . Anantenna gain o . dBi is noticed.Figure depicts the

11


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(c)

F : Field patterns. (a) Open, (b) , and (c) short.

Exterior

Center

F : CRA modication.

2.352.2 2.25 2.3 2.4 2.45 2.5 2.55 2.6

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()

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(g)

F : Field patterns versus microstrip line widths = 90.

: Antenna gain substrate thickness analysis.

Gain (dBi)

= / 6 .

= / 7 .

= / 8 .

= / 9 .

= /10 .

= /11 .

= /12 .

: Optimized antenna dimensions.

Optimized Original

otal diameter . .

Interior arm . .

Exterior arm . .

runcated arm . .

Line width () . .

Angle between interior arms () . .

Angle between exterior arms () . .

Angle interior-exterior arms () . .


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2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6

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h1h2h3h4

h5

h6h7h8h9

h10

F : Return losses or different substrate thicknesses.

F : Cross rhombic antenna with sofened edges.

2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6Frequency (GHz)

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e

f

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F : Return losses or different substrate thicknesses.

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F : Field pattern or = /8.

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F : Field pattern or = /9.

parameter response comparison, with and without sofening,or = /9; as it is seen, there are a higher bandwidthand a better response or the optimized antenna. As a wayo comparison with other kind o antennas, the rectangularmicrostrip slot patch presented in [] has narrower band-width and less gain.

7. Construction and Comparison

wo different antennas were constructed, one o them withsharp edges andthe other with sofened edges. Both antennasare shown inFigure , using RFA substrate with

= 6.15

( = 5 cm at . GHz) and thicknesses o5

= /9.Measurements were done with an Anritsu MSB

Vector Network Measurement System.Figure shows 11

comparison between simulation and construction at 5

=/9 (real) and

7 = /11(optimal); as seen, there is very


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(dB)

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Optimized

F : Comparison between sharp and sofened ends antennas.

(a) (b)

F : Constructed antennas (a) sharp edges and (b) sof edges.

2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6 2.65

Measuredhoptimal

hreal

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(dB)

Frequency (GHz)

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50

0

F : Return losses (antenna with sharp edges).


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2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6

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losses

(dB)

Frequency (GHz)

h4 (measured)

h5(simulated)h4(simulated)

0

5

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25

F : Return losses (antenna with sofened edges).

2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6

Measured

Optimized simulated

Return

losses

(dB)

Frequency (GHz)

0

5

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20

25

30

F : Measured return losses (CRA with sofened edges).

good similarity between them but with a deviation o about MHz over design requency.

On the other hand,Figure shows comparison o11parameters or same antennas but with sofened edges; as it isseen, there is a good similitude between all o them. Finally,Figure depicts coupling comparison between simulationand constructed sofened edges antennas. We see again adeviation o about MHz between both antennas due toconstruction problems. Field patterns or constructed andsimulation antennas are shown inFigure , also with somedifferences.

Figure shows gain comparison o both antennas, andas seen, there are some differences, with a better responseor the simulated one. It is clear that we have to improve ourconstruction methods.

5

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Simulated eld pattern

0

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2530

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0

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6

8

10

2.2 2.25 2.3 2.35 2.4 2.45 2.5 2.55 2.6

Measured

Simulated

Gain(dB)

Frequency (GHz)

F : Gain comparison.

o characterize circular polarization, we obtained axial

ratio, measuring horizontal and vertical eld patterns. Bothpatterns have almost the same magnitude, except in

and , probing circular polarization. Figure presentscircular polarization measurements; (a) shows measured eldpatterns and (b) shows axial ratio obtained rom subtractionmagnitude o both eld patterns.

8. Conclusion

We have presented parametric analysis o CRA with results ogain, eld pattern, and return losses. Results shown o com-paring simulation and constructed structures using RFAwith

= 6.15 agree with most o measured parameters


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50

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(a)

0

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0 50 100 150 200 250 300 350

AR

Axialratio

(deg)

(b)

F : Circular polarization (a) eld pattern comparison. (b) Axial ratio.

although with a displacement o about MHz, meaningthat error in the design requency is due to manuacturinginaccuracy.

As it is seen rom simulation, best conguration is thatusing open load, getting thebest coupling and gain (. dBi inthe simulation and . dBi in measurement). We also analyze

the substrate thickness effects on resonant requency.A better similarity is obtained or the antenna withsofening sharp edges considering its less dependence orequency. We concluded that the best geometry or CRA isthe one with sof edges with better perormance and higherbandwidth.

We have proposed CRA or other applications, as thosepresented in [, ], where we designed and constructedair dielectric antennas or GNSS at . GHz and also anair dielectric antenna or . GHz with . dBi gain andbandwidtharound MHz, orboth o them, are greater thanthose o rectangular or circular patches presented in [,].

Te nal dimensions o antenna are 10 10 0.56 cm

with weight o g, and an excess o . cm over the . cmo maximum antenna dimension or the ground plane. Tose cm per side perectly t in the area o Sensat cube satel-lite.

Conflict of Interests

Te authors declare that there is no conict o interests.

Acknowledgments

Te authors wish to thank to Instituto Politecnico Nacionaland Consejo Nacional de Ciencia y ecnologa de Mexico ortheir economic support.

References

[] J. S. Pedroza, F. M. Zuniga, and M. E. Aguilar, Planar antennasor satellite communications, in Satellite Communications, N.Diodato, Ed., pp. , Sciyo, .

[] A. G. Roederer, Te crossantenna: a new low-prole circularlypolarized radiator,IEEE Transactions on Antennas and Propa-

gation, vol. , no. , pp. , .

[] E. G. Nolasco,Optimizacion de una antena plana para sistemasmultiestandar [M.S. thesis], Seccion de Estudios de Posgradoe Investigacion, Escuela Superior de Ingeniera Mecanica yElectrica Zacatenco, IPN, Mexico City, Mexico, .

[] G. Kumar and K. Ray, Broadband Microstrip Antennas, ArtechHouse, Norwood, Mass, USA, .

[] M..Ali, N.Ramli,M. K.M. Salleh,andM. N.M. an, A designo recongurable rectangular microstrip slot patch antennas,inProceedings of the IEEE International Conference on SystemEngineering and Technology (ICSET ), pp. , Shah Alam,Malaysia, June .

[] J. S. Pedroza, L. E. C. Rivera, S. Pena Ruiz, and F. M. Zuniga,

Analisis de acoplamiento mutuo en arreglos con antenas decruz rombica, in VI Conferencia Internacional en IngenieraElectromecanica y de Sistemas (CIIES ), November .

[] L. C. Rivera, J. S. Pedroza, S. P. Ruiz, and M. A. Mosqueda, Latecnica del patron del elemento activo para analizar los eectosmutuos en un arreglo lineal de antenas Rombicas de Cruz, inXXII Reunion Internacional de Otono,de Comunicaciones, Com-putacion, Electronica, Automatizacion, Robotica y AplicacionesIndustriales y Exposicion Industrial, IEEE, Guerrero, Mexico,December .

[] H.-M. Chen, Y.-K. Wang, Y.-F. Lin, C.-Y. Lin, and S.-C. Pan,Microstrip-ed circularly polarized square-ring patch antennaor GPS applications, IEEE Transactions on Antennas andPropagation, vol. , no. , pp. , .


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[] H. Li, J. Li, D.Li, andY.Zhang,High-gain circular polarizationantenna or small satellite data link application, in Proceedingsof the IEEE International Conference on Signal Processing,Communications and Computing (ICSPCC ), Xian, China,September .


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Research ArticleNovel Dual Band Microstrip Circular Patch AntennasLoaded with ENG and MNG Metamaterials

Md. Ababil Hossain,1 Md. Saimoom Ferdous,2

Shah Mahmud Hasan Chowdhury,1 and Md. Abdul Matin1

Department of Electrical and Electronic Engineering, BUET, Dhaka , Bangladesh Department of ECE, University of British Columbia, Okanagan, BC, Canada

Correspondence should be addressed to Md. Ababil Hossain; [email protected]

Received September ; Accepted November ; Published February

Academic Editor: Guo Qing Luo

Copyright Md. Ababil Hossain et al. Tis is an open access article distributed under the Creative Commons AttributionLicense, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properlycited.

Novel design o a dual band microstrip circular patch antenna loaded with ENG ( negative) metamaterial has been shown inthe rst section. Using ENG metamaterial instead o the conventional dielectric, unconventional M10 (1 < < 2) mode wasproduced to yield a dual band perormance. Optimized parameters such as permittivity (1) and lling ratio () o metamaterialswere selected with the aid o a MALAB based parameter optimization algorithm, developed or all these sort o patch antennas.In the second section, a dual band circular patch antenna loaded with MNG (

negative) metamaterial has been reported. An

unconventional modied M10(0 < < 1) mode has been produced along with conventional M110mode due to using MNGmetamaterial. Here also the optimum values o permeability (1) and lling ratio () or these sorts o patch antennas have beencalculated rom a MALAB based parameter optimization algorithm. Both the proposed antennas provide good and resonanceand satisactory radiation perormances (directivity, radiation efficiency, and gain) with a dual band perormance.

1. Introduction

Satellite communications, wireless communications, surveil-lance,weather, radar, andso orth require highly directiveand

multiband antennas. In act, in recent times, highly directivemultiband antennas have become a prime concern in satellitecommunication.

Te use o multiband antenna has brought about revo-lutionary change and dynamism in the trends o commu-nication over the last ew decades. However, antenna sizeminiaturization is also an important actor as ar as antennasize is concerned. Tereore, a unique radiating element suchas patch antenna which incorporates multiband perormancewith size miniaturization can urther open up a new horizonin telecommunication sector. In [], a generalized MALABbased parameter optimization algorithm or designing cir-cular patch antenna loaded with metamaterials has been

developed. On the basis o that design algorithm,in thispaperwe have presented the design model or dual band antennasloaded with both ENG and MNG metamaterials without theuse o any symmetrical slotting which was done in [,].

Wong and Hsieh [] constructed dual band circular patchantenna by using conventional dielectrics as substrate andby using symmetrical slots where conventional M210modewas modied to yield dual band perormance. In that casethe rst order mode (M110 mode requency) and secondorder mode (M210mode requency)were determined by theantenna geometry structure (i.e., radius o the patch). Oncethe one mode has been selected, the other mode requencyautomatically becomes xed. In that case the designer couldonly tune any one mode requency according to his will;he had no control over choosing the other mode requencyas the other mode requency is determined by antennaradius. In that particular case the degree o reedom was


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only one, that is, radius o the patch. So with one degree oreedom designers could choose only one resonant requencyindependently, but not both at the same time.

In the rst section o this paper, a novel design odual band ENG metamaterial loaded circular patch antennahas been shown. Its main eature is in the exibility o

choosing both resonant requencies according to users willand apart rom conventional M110 mode newly producedunconventional M10 (1 < < 2) mode provides highdirectivity. Our designed antenna has two degrees o reedom(radius o the patch and lling ratio o the metamaterial). Inourcase, the rst band, that is,rst ordermode, is determinedby the patch geometry only. Using ENG metamaterial insteado conventional dielectrics causes resonance in between rstandsecond ordermode requency. For thisreason, thebiggestadvantage here is that the designer can tune the secondband almost anywhere in between conventional rst andsecond order mode by just changing the lling ratio o themetamaterial. Tat is why this design provides a great deal o

exibility in choosing resonant requency in comparison toother dual band circular patch antennas reported so ar.Alu et al. proposed design method to obtain electrically

small rectangular patch antennas usingDPS (double positive)ENG metamaterial juxtaposed layer []. But such rectangularpatches give broadside null radiation pattern at subwave-length regime as illustrated in Figure o []. Finally, it waspredicted that all these electrically small rectangular antennasloaded with metamaterial can only be good resonators butmay not be good radiators. But i the shape is circular[,] or elliptical [], theoretically it is possible to achieveelectrically small size without deterioration o radiationperormance. With this concept we developed a properalgorithm in [], where achieving additional unconventionalmode in metamaterial loaded circular patch antenna hasbeen possible. By using the same algorithm, here a dualband ENG metamaterial loaded patch antenna has beendesigned. Highly directive perormance and exibility intuning requency are the two prime criteria o this dual bandpatch antenna.

In the second section o this paper, by using MNG (negative) metamaterial in the inner core as substrate insteado natural dielectric in circular microstrip patch antenna,dual band perormance has been achieved with a reducedsize antenna. Apart rom conventional M110 mode, hereunconventional M10 (0 < < 1) mode has beenproduced. Applying our proposed design algorithm [], wecan achieve broadside radiation at M10(0 < < 1) modeand hence achieve a dual band perormance with a reducedsize antenna. In this sort o metamaterial loaded patchantennas, the resonant requency can be tuned accordingto designers will. But it has been shown that electricallysmall size patch antenna having metamaterial loading is notpracticable without degradation o radiation perormance[, ]. In act, the gain perormances o such electrically smallantennas degrade rapidly due to the size reduction. However,nearly% size reductionmay be possible where gain remainsabove dB []. In our proposed designed antenna based onour developed algorithm, we have obtained around % sizereduction with a gain o . dB. Flexibility in tuning resonant

requency along with size miniaturization has certainly madethis dual band patch antenna unique rom other dual bandantennas reported in current literature.

MNG metamaterial can be abricated in the laboratory atmicrowave requencies by using Split Ring Resonators (SRR)and helices, whereas thin wired structure placed at a periodic

distance works as negative Epsilon material when the dis-tance between consecutive wires becomes comparable to thewavelength o the propagating electromagnetic wave. Overthe years, a lot o research has been carried out in ENG loadedpatch antenna, but ironically the practical implementation oENGmetamaterial at microwave regimeusing Lorentz modelis a tough task []. However, recent advancement o ENGmedium applications in microwave regime urges its practicalimplementation. Although the practical application o ENGmedium using Lorentz model in microwave regime is reallya tough one, the directivity and gain perormance o suchantennas are attractive with conventional size.

2. Mathematical Modeling of MetamaterialLoaded Circular Microstrip Patch Antenna

Te basic magnetization vector equation or a circular patchantenna can be written as

, , = + cos + sin 3cos + 3sin .

()

But radial component o the electric eld can be derived as

= 1 2

. ()Here, = 1or region (metamaterial region) and = 2orregion (conventional dielectric region).

From boundary conditions (perect electric wall assump-tion o cavity model),

at = ,= 0, we get3= 0,and at = 0,= 0, then we get3= 1.

Tereore,

, , = + cos + sin cos ,

()

, , = + cos + sin sin .

()


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Generalized equation or the phi component o the electriceld is as ollows:

= 11

2

. ()By calculation rom the equation o

we get

,, = + sin + cos sin .

()

Finally,-component o electric eld becomes=

2

2+ 2. ()

Again, calculation rom the equation oyields, , =

2 +

cos + sin cos .()

For the magnetic elds component o the cavity,-compo-nent o the magnetic eld is everywhere.

So,

= 0. ()Again, radial component o the magnetic eld is ormulatedas

= 1

= + sin + cos cos .

()

Finally,

= 1, ()

which gives

= +

cos + sin cos .()

For all the above equations, and are Bessel unctionso rst kind and its derivative denoted by prime order o

. and are Bessel unctions o second kind and itsderivative denoted by prime order o.

Now applying boundary condition at the interace, that is,

at = , 1= 2 we get rom electric eld equations,

2111 11

2222 22

222222= 0.()

Again, at = , 1= 2, we get rom magnetic eldequations that

11 11 22

22

22

2

2

= 0.

()

Finally, at = , 2= 0 and then we get similarly22

22+ 22

22= 0. ()

From the previous three equations, that is, (), (),and (), we see that the right side is zero. So, in order tohave solutions, the determinant ormed by the correspondingmatrix must be equal to :

211

1

1 22

2

2

2 22

2

2

211 1

22 2 22 2

0 22 2 22 2

= 0.

()

Solving the above determinant we get

1111

=22 2 2 2 22 2 22 .

()

Now or ENG metamaterial,

1 1= 1 1. ()And or MNG metamaterial,

1

1

= 11.

()


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ENG1

h

tg y

z

xa

DPS2

Dg

Region 1

Region 2

a

F : Geometry o a circular microstrip patch antenna partiallyloaded with metamaterial (ENG) substrate and ground plane with

parameters = (.56) (20mm) = 11.2mm, = 20mm, =4.3mm,= 2mm,= 40mm,2= 1.33,1= 1.9 (at. GHz),2= 1,1= 1, and eed position rom the centre= 15mm. : Selected permittivity and lling ratio or ENG-metamate-rial loading in antenna.

Chosen resonantrequency,(GHz) 1(at) Filling ratio (). . .

Putting the above relation, the dispersion relation orENG metamaterial loaded patch antenna becomes

2122 1111 + +11

=1 1 1 12 2 2 2 .()

And or MNG metamaterial, the dispersion relation is

2 1 221 111 + +11

=11 1122 22 .()

By using the above two dispersion relations, a MALABbased parameter optimization algorithm has been developedin []. In that algorithm, the optimized value o metamaterialparameter such as lling ratio, permittivity, and permeabilityhas been calculated or metamaterial loaded circular patchantennas.

30

20

10

0

10

20

3.5 4 4.5 5

Frequency (GHz)

Lorentz model of ENG

Re[]

Im[]

F : Lorentz dispersive model o ENG or proposed antenna.

0

5

10

15

20

Frequency (GHz)

2 3 4 5

S11 parameter (dB)

F :11 parameter or mm circular patch antenna usingENG as core material.

3. ENG Metamaterial Loaded Dual BandCircular Patch Antenna

.. Antenna Design Structure and Specications. In circularmicrostrip patch antenna,by using ENGas a substrate insteado regular dielectric we can achieve dual band perormance.Apart rom conventionalM110 mode unconventional M10(1 < < 2) mode can be modied.

In designing our proposed antenna, geometry o a cir-cular patch with metamaterial block as shown in Figure has been used. With proper choice o lling ratio(),ENG metamaterial(1)is loaded concentrically with regulardielectric(2). In Figure ENG metamaterial is denoted byregion (green) and regular dielectric is denoted by region (yellow). Te antenna consists o a metallic patch o thickness(= 1mm) and radius ( = 20mm) at the top. A metallicplate placed at the bottom with radius (

2 = 40mm) and


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(a) (b)

F : D view o (a) conventional M110mode at . GHz and (b) unconventional M10(1 < < 2) modes radiation pattern at. GHz.

(a) (b)

F : Electric eld distribution on the = 0plane at (a)110= 3.6665GHz and (b)10(1 < < 2)=. GHz.

thickness (= 2mm) acts as ground. Tis chosen optimumradius and thickness o the ground plate causes maximumreection rom the ground that enhances the directivity.Conductivity o the metal is selected as . Meg. As a eedwe have used coaxial cable designed to have= 50characteristic impedance with inner radius o coaxial cable(in= 0.4mm) and outer radius o coaxial cable (out=1.0456mm) and its position is set at= 15mm rom thecenter or achieving good matching properties overoperatingrequency range. Te underneath substrate is lled withconcentric dielectrics with ENG metamaterial as the innercore and DPS material as outer core. Dielectric substrateheight is = 4.3mm. O all the dielectric parameters,2=1.30,2= 1.00, and1= 1.00are chosen as the optimum

values or these parameters. However, the rest two controllingparameters, that is, ENG metamaterials permittivity (

1) and

lling ratio () have been calculated by a MALAB basedparameter optimization algorithm [] and with the aid odispersive equation (), developed orENG metamaterial. Inable. those optimum parameters are shown at our desiredresonant requency o . GHz.

Actually, SNG or DNG metamaterials are inherentlydispersive and lossy [, ]. So, without using dispersivelossy model the simulated results cannot give proper realisticresults. Here we have used Lorentz model or ENG metama-terial dispersive relation:

()= + 20

20

2

, ()

where= 1.00,= 1.230,0= 26.6Grad/sec,=4.26GHz, and damping requency = 1MHz.Plugging these parameters into () dispersive relation o

our desired ENG material should be as in Figure.Using all the above stated material and geometric param-

eters, CS microwave studio [] simulation gives the ollow-ing results:-parameter, D radiation patterns, and electricand current distribution (Figures).

.. Resonance and Radiation Characteristics of the Antenna.From theS11parameter curve o we see that correspondingreturn losses at . GHz and . GHz requencies allwell below dB that ensures satisactory resonance orthe antenna structure. But satisactory resonance does notalways guarantee good radiation. So in order to ensuregood radiation and thus call it a good radiator, we need tolook into its radiation patterns at those particular resonantrequencies also. Te radiation patterns obtained at theresonant requencies indicated at the11parameter curve oFigureare shown in Figure.

From the D radiation patterns o the ENG loaded patchantenna (see Figure), we seethatconventional M110modeas well as the modied M10 mode (1 < < 2) showssatisactory-directed radiation (Figures(a)and(b)). Teconventional M110 mode has the gain o . dB, whereasthe gain o the modied mode M10 (1 < < 2) atrequency . GHz is . dB, which is even higher thanthe conventional mode.


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(a) (b)

F : Current distribution on the patch at (a)110= 3.6665GHz and (b)10(1 < < 2) = 4.2686 GHz.

MNG1 DPS2

h

tg y

z

xa

Dg

Region 1

Region 2

a

F : Geometry o a circular microstrip patch antenna partiallyloaded with metamaterial (MNG) substrate and ground plane withparameters: = (.56mm)(20mm) = 11.2mm, = 20 mm, =5mm,= 2mm,= 40mm,2= 1.33,1= 1.33,2= 1,1= 2.83(at . GHz), and eed position= 15mm.

.. Electric Field and Current Distribution. From the electriceld distribution at the plane = 0, it is apparent that incase o conventional M110 mode electric eld ips its signpassing rom one side to the other side o the patch, thussatisying the condition or broadside radiation (Figure (a)).But in case o modied M10(1 < < 2) mode due to usingmetamaterial, phase change o electric eld at the ENG-DPSinterace occurs in such a way that the electric eld ips romone side to the other side o the patch, as a result o which theelectric eld distribution o M10(1 < < 2) mode lookslike that o M110mode, thus also satisying the condition orbroadside radiation (Figure(b)).

Microstrip circular patch antenna theory says that i themetallic patch induced current remains in the same phasepassing rom one side to the other side o the patch, thatis, current distribution remains symmetric around-axis, itmust show broadside radiation. At ( = 0, = 0, = 0) pointmagnetic eld distribution rom the two opposite current

Lorentz model of MNG

Frequency (GHz)

20

10

0

10

20

1 1.5 2 2.5 3 3.5 4

Re[]Im[]

F : Lorentz dispersive model or proposed MNG loadedantenna.

elements reinorce in direction. Tat is why M110 modeshows -directed radiation (Figure (a)). But any other modedoes not normally show this property. What metamaterialdoes is that, in our desired requency, at interace o DPS-ENG, it causes phase change in such a way that the currentdistribution at that requency resembles with M110 modes

current distribution. Figure(b)shows current distributiono the patch or M10(1 < < 2) mode that resembles withM110modes pattern. It is noticeable that, despite the smalldimension o the patch, the current distribution is closed inelectrically small resonant loops and it resembles the currentdistribution o M110mode.

4. MNG Metamaterial Loaded Dual BandCircular Patch Antenna

.. Antenna DesignStructure andSpecications. In designingour proposed antenna, geometry o a circular patch withmetamaterial block as shown in Figure has been used. It


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0 1 2 3 4 5

0

10

30

20

10

Frequency (GHz)

S11 parameter (dB)

F :11-parameter or mm radius circular patch antennausing MNG as core material.

is almost similar to the ENG loaded patch structure. It alsoconsists o a metallic patch o thickness (

= 1mm) and

radius ( = 20mm) at the top. A metallic plate placed atthe bottom with radius (2 = 40mm) and thickness(= 1mm) acts as ground. Tis chosen optimum radiusand thickness o the ground plate causes maximum reectionrom the ground that enhances the directivity. Te metalhas a conductivity o . Meg. As a eed we have usedcoaxial cable designed to have= 50 characteristicimpedance with inner radius o coaxial cable (in= 0.4mm)and outer radius o coaxial cable (out= 1.0351mm) and itsposition is set at= 12mm rom the center or achievinggood matching properties over operating requency range.Te underneath substrate is lled with concentric dielectricswith MNG metamaterial as the inner core and DPS material

as outer core. Dielectric substrate height ( = 5mm) isspacious enough so as to allow realistic hosting o split ringresonator (SRR) to construct MNG material []. O all thedielectric parameters,2= 1.30,2= 1.00, and1=.0 are chosen as the optimum values or these parameters.However, MNG metamaterials permeability (1) and llingratio () have been calculated by a MALAB based parameteroptimization algorithm [] and with the aid o our deriveddispersive equation () or MNG metamaterial. Calculatedoptimum permeability and lling ratio are shown in able.

We have used Lorentz model or MNG metamaterialdispersive relation:

()= + 2

020 2 , ()where= 1.00,= 1.50, and0= 14.76Grad/seccauses resonance at= 2.56GHz and damping requency( = 10MHz). Putting these values in (), we get the disper-sion curve or MNG as shown in Figure .

.. Resonance and Radiation Characteristics. Using allthe above stated material and geometric parameters, CSmicrowave studio [] simulation gives the ollowing results:-parameter (Figure), D radiation patterns, and electriceld distribution (Figuresand).

: Selected permeability (1) and lling ratio () or MNGloading.

Chosen resonantrequency,(GHz) 1(at) Filling ratio (). . .

11 parameter curve, in Figure, shows that at requen-cies10(0 < < 1) = 2.56GHz,110= 3.8GHz resonanceoccurs, which causes the corresponding return losses at theserequencies to all well below dB that ensures satisactoryresonance. Again, rom Figure, it is also evident that whenantenna size is reduced, resonant requency has to move awayrom the M110 requency which eventually causes to loseits-directed radiation characteristics. So it is not alwaysproductive to reduce patch antenna size whimsically. But, inour study, we have ound that around % size reductionprovides us with nearly dB gain, whereas size reduction,that is,

110 10(0 < < 1)110 100%=(3.66GHz 2.56GHz)3.66GHz 100% 30%,

()

providesus . dB gain which is quite satisactoryor practicalantenna operation. Tis is the maximum allowable limit osize reduction with realizable gain perormance that we haveound in our study. In act, antenna size reduction can beextended up to nearly % to get gain over dB.

From the D radiation patterns o MNG loaded antenna(see Figure) we see that conventional M110mode as well

as the modied M10 mode (0 < < 1) shows satisactory-directed radiation. Te conventional mode M110has thegain .dB (Figure(b)) at requency . GHz, whereas thegain o the modied mode M10(0 < < 1) at requency. GHz is . dB (Figure(a)), which is much lower thanthe conventional mode. So, this sort o dual band antenna canbe a suitable choice where low gain and less directive dualband perormance is a prior requirement.

.. Electric Field Distribution. From the electric eld distri-bution at the plane = 0, it is apparent that or conventionalM110mode electric eld ips its sign passing rom one sideto the other side o the patch, thus satisying the condition

or broadside radiation (Figure (b)). But in case omodiedM10(0 < < 1) mode due to using metamaterial, phasechange o electric eld at the MNG-DPS interace occurs insuch a way that the electric eld ips rom one side to theother side o the patch, as a result o which the electric elddistribution o M10 (0 < < 1) mode looks like that oM110mode, thus also satisying the condition or broadsideradiation (Figure(a)).

5. Conclusions

In the rst section o this paper, a novel dual band cir-cular patch antenna with high directivity perormance in


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(a) (b)

F : D view o (a) unconventional M10 (0 < < 1) modes radiation pattern at . GHz and (b) conventional M110 modesradiation pattern at . GHz.

(a) (b)

F : Electric eld distribution on the = 0plane on the metallic patch or the antenna o Figureat (a) . GHz or unconventionalM10(0 < < 1) mode and (b) . GHz or conventional M110mode.

both bands has been shown. Inserting ENG metamaterialprovides us with an additional modied mode with highgain. By controlling ENG metamaterials lling ratio differentrequency can be tuned in. So, this novel antenna designcan be a suitable choice where highly directive gain alongwith users exibility in choosing resonant requencies ordual band antenna purpose is required. In the second sectiono this paper, a miniaturized size dual band antenna loadedwith MNG metamaterial has been designed. In this sort oantenna, since the unconventional mode yields a much lowergain than the conventional mode, one o its dual bands can beused as a substitute or low-gain antenna. Actually, this sorto antenna can be very effective where theuse o less directiveand low-gain small antennas is a prime concern. Te use othis MNG loaded patch antenna is expected to be conduciveor reasonably well signal propagation in some other specialareas, such as high multipath intererence regions, regionshaving requent obstacles between transmitters and receivers,highly dense populated areas, and in spacecraf as a backup tothe high-gain antenna.


Te authors declare that there is no conict o interestsregarding the publication o this paper.

References[] S. Ferdous, A. Hossain, S. M. H. Chowdhury, M. R. C.

Mahdy, and M. Abdul, Reduced and conventional size multi-band circular patch antennas loaded with metamaterials,IETMicrowaves, Antennas & Propagation, vol. , no. , pp. ,.

[] K.-L. Wong and G.-B. Hsieh, Dual-requency circularmicrostrip antenna with a pair o arc-shaped slots,Microwaveand Optical Technology Letters, vol. , no. , pp. , .

[] A. Alu, F. Bilotti, N. Engheta, and L. Vegni, Subwavelength,compact, resonant patch antennas loaded with metamaterials,IEEE Transactions on Antennas and Propagation, vol. , no. ,pp. , .

[] K. S. Zheng, W. Y. am,and D.B. Ge, Broadside subwavelengthmicrostrip antennas partially loaded with metamaterials, inProceedings of the International Workshop on Metamaterials(META ), pp. , Nanjing, China, November .

[] F. Bilotti, A. Alu, and L. Vegni, Design o miniaturizedmetamaterial patch antennas with-negative loading, IEEETransactions on Antennas and Propagation, vol. , no. , pp., .

[] P. Y. Chen and A. Alu, Sub-wavelength elliptical patch antennaloaded with-negative metamaterials,IEEE Transactions onAntennas and Propagation, vol. , no. , pp. , .

[] J. Pruitt and D. Strickland, Experimental exploration o meta-material substrate design or an electrically small patch-like


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antenna, in Proceedingsof the Antennas and PropagationSocietyInternational Symposium (APSURSI ), IEEE, July .

[] S. Jahani, J. Rashed-Mohassel, and M. Shahabadi, Miniatur-ization o circular patch antennas using MNG metamaterials,IEEE Antennas and Wireless PropagationLetters, vol. ,pp., .

[] J. Xiong, H. Li, B. Z. Wang, Y. Jin, and S. He, Teoreticalinvestigation o rectangular patch antenna miniaturizationbased on the DPS-ENG bi-layer super-slow M wave,Progressin Electromagnetics Research, vol. , pp. , .

[] M. R. C. Mahdy, M. R. A. Zuboraj, A. A. N. Ovi, and M. A.Matin, An idea o additional modied modes in rectangularpatch antennas loaded with metamaterial,IEEE Antennas andWireless Propagation Letters, vol. , pp. , .

[] M. Hassan, M. R. C. Mahdy, G. M. Hasan, and L. Akter,A novelminiaturized triple-band antenna, in Proceedings ofthe th International Conference on Electrical and ComputerEngineering (ICECE ), pp. , Dhaka, Bangladesh,December .

[] CS Microwave Studio, CS o America,http://ww w.cst.com.

[] S. M. H. Chowdhury, M. A. Hossain, M. S. Ferdous, M. R.Chowdhury Mahdy, and M. A. Matin, Conceptual and prac-tical realizationo reduced size multi-band circular microstrippatch antenna loaded with MNGmetamaterial, in Proceedingsof the th International Conference on Electrical and ComputerEngineering (ICECE ), pp. , Dhaka, Bangladesh,December .


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Research ArticleMultishorting Pins PIFA Design for Multiband Communications

Muhammad Sajjad Ahmad, C. Y. Kim, and J. G. Park

School o Electronics Engineering, Kyungpook National University, Sankyuk-Dong, Buk-Gu, Daegu -, Republic o Korea

Correspondence should be addressed to C. Y. Kim; [email protected]

Received October ; Revised December ; Accepted December ; Published February

Academic Editor: Bing Liu

Copyright Muhammad Sajjad Ahmad et al. Tis is an open access article distributed under the Creative CommonsAttribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work isproperly cited.

A novel PIFA model with multiple shorting pins is proposed or multiband, low prole wireless applications, which has the abilityto work in adverse conditions. Te proposed model has a planar radiating sheet, a ground plane, and sides covered with PECboundaries. Te substrate inside the antenna box is tempered in order to improve the bandwidth and gain. Te enhancementsapplied to the proposed PIFA model show improved characteristics or this PIFA model and make it a versatile candidate orhandheld, low prole, and multiband resonant communication devices. Pertinent communication devices are those that work withGSM /, UMS ////, LE /, and ISM bands used or Bluetooth and WLAN.

1. Introduction

Antennas (electromagnetic waves guiding devices) radiatesignals to unbounded mediums. Tey are requency depen-dent devices and are designed to operate at specic requen-cies known as the antennas operatingbands. Other than thesespecic requency bands, an antenna rejects any signal thatis ed to it. Antennas are known or their various propertiesincluding gain, directivity, radiation pattern, specic absorp-tion rate (SAR), and Voltage Standing Wave Ratio (VSWR).

In [] modied planar inverted-F antenna (PIFA)models were proposed with compact size, multiple resonant

bands, and enhanced bandwidth by changing the widtho the shorting and eed pins, adding a parasitic elementparallel to the shorting pin at an optimized distance and aplanar rectangular monopole top loadedwith two rectangularpatches with one o them grounded, respectively. In anordinary PIFA model, when a shorting pin is applied nearthe eeding point, it allows the design to be reduced in sizebut narrows the bandwidth at the same time. By applyingdifferent schemes and techniques to an ordinary PIFA model,we can enhance not only its bandwidth but also its gain andefficiency as well.

In recent years, the antenna industry has shown a rapiddemand or multiband resonant, low prole, and ultrawide

bandwidth antennas []. Te act that PIFA has a exibledesign and can provide multiband resonantoperations makesit a avorable candidate or the antenna industry. Introducingslots in the radiating patch may allow designers to achieveresonant requencies that are not possible or a conventionalPIFA with small dimensions to resonate on. Moreover, withslots in the ground plane o a PIFA model, it is reportedin the literature as a bandwidth enhancement method [].Such PIFA models, commonly known as a meanderedgroundplane or meandered radiating patch, have diversied PIFAsor the low prole design industry. Plenty o modied PIFAmodels have been designed or multiband operations; slottedPIFA, meandered ground plane PIFA, and multiple shortingpins PIFA have been successul among others. In [, ],the limitations o electrically small antennas in terms o thegain, bandwidth, and their capacity to produce the desirednumber o resonant bands along with guidelines to designsuch antennas based on the parametric analysis o electricallysmall antennas are discussed.

Antennas are designed andtested in almostideal environ-ments, but when they are exposed to conducting materialsin their surroundings, they do not just shif their reso-nant requency but the bandwidth and gain are changedas well. Te perormance o any conventional antenna isaffected severely, in terms o its resonant requency, gain,


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Antenna

PEC case

box

135 mm

65mm

15 mm

30mm

F : Proposed PIFA model antenna boxand PEC case, withPEC boundaries around it.

and bandwidth, in the presence o conducting bodies aroundthe antenna. It is considered a loophole or communicationdevices working with such antennas. In [, ], the impacto metallic suraces and users hand on the perormance odifferent antennasare discussed. Te efficiency o the antennais reported to drop rom % (without the hands effect) to% (with the hands effect), which might not be acceptableor communication devices which require higher efficiency.Tereore, an exquisite antenna design is needed or todaysindustry which can cope with such undesired situations andmaintain its efficiency even in the worst conditions. Ourproposed model has the ability to maintain its perormancein such critical conditions. Details o our proposed antennadesign are given in the ollowing section.

2. Antenna Design ProcedureTe proposed antenna model is shown in Figure. It consistso a dielectricmaterialFR-, which hasa dielectricconstant o= . and a thickness o mm,antenna box, andPEC case.Te dielectric material is sandwiched between the radiatingpatch (top sheet o the antenna box) and the ground planeo thePEC case. Te dimensions o the proposed model are mm3. For simplicity the model is divided in twomain sections: rstly, the PEC caseand second theantenna

box. Te dimensions o theantenna boxare mm3.Te top sheet oantenna boxacts as a radiating patch whichcontains slots around itsboundary to maintain the separationbetween the antenna elements and thePEC case. In Figures and, a D view o the proposed model is shown withspecial emphasis on the antenna box and its components,that is, the radiating patch, parasitic patch, eeding position,and shorting pins. A parametric description or the radiationpatch is presented in Figure, with detail o the parametersin able. Slots in the radiating patch are used to increase theelectrical length o the radiating patch in order to achieve theresonant band at lower requencies. Te sides o theantennaboxare acting as parasitic patch elements. A prototype o theproposed model is designed and shown in Figure.

Te proposed model contains three shorting pins, shownin Figure, that are connecting the radiating patch to theground plane at different point. Te width o each shorting

: Length o the slots inserted in the radiating patch.

Slot name Length (mm)

A

B

A

B 1 .

2

3

4

5

6

7

8

pin is mm and the height is mm. Tickness o the model

is mm. Te width o the lumped port eeding sheet is mmand the direction o integration line or the modes excitationisalongthe -axis. Te distance between consecutive shortingpins can be changed to shif the resonant bands i desired.Multiple shorting pins and a shortening pin are used toadd stability to this model and to obtain multiple resonantbands [,]. Because the boundaries are PEC and the PIFAantenna models are known or their narrow band operations,we used the two sides o the antenna box as a parasiticpatch. In [,], the parasitic patch or microstrip antennaswas introduced and a bandwidth enhancement o .% wasreported. Itwas established thatmultiple parasitic patches canbe used or bandwidth enhancement purposes, both in the

vertical and horizontal positions. In our model, the parasiticpatch is perpendicular to the driving patch and is connectedto the ground plane through the shortening pin.

.. Slots in the Radiating Patch. Slots are inserted in theradiating patch to increase the electrical length o the radi-ating patch. Te width o the slots in the radiating patch is. mm and is denoted by; the slot lengths were differentand are given in able . Te slots around the boundaryo radiating patch, that is, A, B, A, and B, are usedto maintain a separation between the antenna elements otheantenna boxand PEC case; their width is xed to mm.Although the width and length o the slots help us to changethe resonant requency bands, the interslot coupling effectcannot be neglected either. Both the distance between theslots and their width allow us to tune our model to a suitablecoupling effect or desired S results.

It is known that the small size o a radiating patch hasa limit or producing resonant bands at lower requencies; awell-tuned slotted model can achieve resonant bands at lowerrequencies and may improve the perormance o the model[]. Using the act that inserting a slot in the radiating patchmay result in a new resonant requency, the position andlength o the slot to be inserted in the radiating patch or adesired resonant requency can be approximately predicted[].


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PEC case

Slots

Shorting pins

Radiating patch

Feeding pin

Parasitic patch

Shortening pin

Antenna box

F : A D view o the model, parasitic patch, radiating patch,slots, and shortening pin.

.. Multiple Shorting Pins. In [], a dual shorting pinPIFA model was proposed and designed or the dual bandoperationso mobilehandsets.Multiple shorting pins providedifferent paths and lengths to the antenna or radiatingmultiple requencies. Shorting pins provide multiple pathsto currents and allow the antenna to radiate at multiplerequencies. In PIFA, the resonant requency bands dependon both the position and width o the shorting pins used inthe model []. Multiple shorting pins add stability to theantenna model by allowing it to maintain its perormancein adverse situations. Shorting pins, when applied near the

eed position, allow designers to reduce the size o themodel. In our model, the positions o the shorting pins werechosen careully to enhance the perormance o the antennaat the desired requency bands and to suppress undesiredrequencies.

.. Parasitic Patch. A parasitic patch is used to control thedirectivity and is useul in many ways specially designinglow prole antennas. Te parasitic patch has a dual effect onS, when used with the PEC and PMC boundaries. Te twosides o the antenna boxshown in Figure are being usedas a parasitic patch to improve the S result o our design.Parasitic patches are widely being used in antennas to change

the radiating eld patterns, steer the beam, and increase thebandwidth []. Te parasitic patch in our model is usedwith PEC boundary. Parasitic patch with the PMC boundarymay act like a high impedance surace (HIS) []. HIS basedantennas are extensively being used in vehicular antennas.

.. empering the Substrate. Dielectric materials like FR-are used in antenna designs or many reasons. One aspect isthat because it allows miniaturization o the antenna model,and at the same time it is very cheap. However, the problemwith using a bulk o dielectric material in communicationdevices is that it effects the perormance o the antenna interms o efficiency. Because o antenna size limitations (as

Shortening pin

Slots

Shortening pin

Parasitic patch Integration line

FeedingRadiation patch

X

YZ

F : A D view o the antenna box, radiating patch, parasiticpatch, eeding, and shorting pins.

Shorting pin A

Shorting pin B

Shorting pin C

Region A Region B Region C

Feed

S

= mm

L4

L3

L2

L1

LA

L7

L6

LB

WB

WA

L8

L5

F : Parametric description o the slots in the radiation patch.

we cannot go on increasing the height o our model), itbecomes an energy storing device which may be reerred toas a lossy device. o deal with this situation, as shown inFigure, our vacuum gaps are inserted in the substrate atdifferent positions and the width has been swept or multiple

values to ultimately make the proposed model radiate themaximum energy with improved efficiency []. Te purposeo an antenna is to radiate a signal that is ed to it and not tostore it and increase the antenna losses. In [], substrates arediscussed, which can be used in antenna designs. Moreover,the effects o different substrates on the perormance oantennas are also highlighted to control the antenna losses.

In our model, we have inserted vacuum gaps insidethe FR- dielectric material (which we reer to as substratetempering), or which both the position and width o the

vacuum gap affect the S results o our model. Moreover, theresults show that tempering the substrate material may alsoimprove the gain in the resonant bands o the antenna [].

Because the proposed antenna model is a kind o cavity,the cavity perturbation method can be applied to it. Anyincrement or decrement in or at any point in the cavity


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Vacuum gaps

Z

X

Y

wg

wg

wgwg

F : Substrate tempering by inserting vacuum gaps in FR-.

S11(dB)

Frequency (MHz)

0

Region ARegion BRegion C

1 2 3

0

5

10

15

20

(a)

S11

1 2 3

0

5

10

15

20

L4 = 20 mmL4 = 22 mmL4 = 24 mm

(b)

S11(dB)

Frequency (MHz)

0.5 1.0 2.0 3.02.51.5

0

5

10

15

25

20

s = 0.5 mms = 0.25 mm

s = 0.75 mm

(c)

S11(dB)

Frequency (MHz)

2 31

0

5

10

15

20

wg = 1 mm

wg = 1.5 mm

wg = 2 mm

(d)

F : (a) Shorting pins position effect on S. (b) Effect o slot length on S. (c) Effect o slot width on S. (d) Effect o varying the widtho vacuum gaps inserted in FR- substrate on S.


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0 1 2 3

0

5

10

15

20

6.25

Simulated result

Measured result

S11(dB)

Frequency (MHz)

F : Comparison o simulated and measured S results or the proposed model.

Feed(a)

(b) (c)

PEC case

PEC case

Antenna box

F : (a) Antenna box andPEC case. (b) op view. (c) Bottom view.

may decrease or increase the resonant requency o the cavity.Moreover, changein resonant requency can also be related tothe stored electric and magnetic energies inside the cavity aswell. We choose the orientation and position or the vacuumgap using parametric sweep optionin D simulation tool usedor designing this model. Considering that the elds insidethe cavity are approximately the same beore and afer thesubstrate tempering or perturbation, we may conclude thatthe resonant requency o the cavity may increase or decreaseafer tempering the substrate depending upon the position otempering or perturbation inside the cavity [,].

3. Results and Discussion

Te proposed model is simulated with the High FrequencyStructural Simulator (HFSSv.) and a prototype or theproposed model is also designed. Te comparison o thesimulated and measured S results is shown in Figure .Section. deals with the benets o the enhancements weapplied to the model. In Section., S and the magnitudeo the E-eld at the corresponding resonant requencies areelaborated. Section. covers the details o the bandwidthsand the gains at corresponding resonant requencies.


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4.1388e 0011.8392e + 0003.2645e + 0004.6898e + 0006.1150e + 0007.5403e + 0008.9656e + 000

1.0391e + 0011.1816e + 0011.3241e + 0011.4667e + 0011.6092e + 001

1.8943e + 0012.0368e + 0012.1793e + 0012.3219e + 001

Z

X

Y

dB (gain total)

1.7517e + 001

(a)

1.8511e + 000

Z

X

Y

dB (gain total)

8.8131e + 0009.4460e + 0001.0079e + 0011.0712e + 0011.1345e + 001

5.6485e + 0006.2814e + 0006.9143e + 0007.5472e + 0008.1801e + 000

2.4840e + 000

1.2182e + 000

3.1169e + 0003.7498e + 0004.3827e + 0005.0156e + 000

(b)

9.0868e 001

Z

X

Y

dB (gain total)

1.1168e + 0011.2101e + 0011.3033e + 0011.3966e + 0011.4898e + 001

6.5046e + 0007.4372e + 0008.3699e + 0009.3025e + 0001.0235e + 001

1.8413e + 000

2.3972e 002

2.7740e + 0003.7066e + 0004.6393e + 0005.5719e + 000

(c)

1.7888e + 000

Z

X

Y

dB (gain total)

1.3199e + 0011.4236e + 0011.5273e + 0011.6310e + 0011.7348e + 001

9.0496e + 0001.0087e + 0011.1124e + 0011.2161e + 001

2.8260e + 000

7.5152e 001

3.8633e + 000

4.9006e + 000

6.9751e + 0008.0123e + 000

5.9378e + 000

(d)

F : (a) Gain at the resonant requency MHz. (b) Gain at the resonant requency . GHz. (c) Gain at the resonant requency. GHz. (d) Gain at the resonant requency . GHz.

.. Effects o the Enhancements. Te enhancements that wehave applied to the proposed model have resulted in allowingus to obtain the wide bandwidths at the resonant requencies.Te enhancements are inserting slots in the radiating patch,a parasitic patch with the PEC boundary, multiple shortingpins, and substrate tempering by inserting vacuum gapsinside the FR- substrate. Te effects o these enhancementson the S curve are clear in Figure.

In Figure (a), the shif in the resonantrequencybands isshown when shorting pins are swept along region A, regionB, and region C as mentioned in Figure . Te S result inFigure (a) clearly shows that the PIFA model provides anarrow bandwidth when the shorting pins are close to theeeding pin. Because the electric length between region A andthe eed point is the shortest compared to regions B and C,the bandwidth is narrow. When shorting pins are applied inregion C, the bandwidth is wider. o increase the bandwidtho our proposed model, the shorting pins can be swept alongregions A, B, and C, respectively.

In Figures(b)and(c), the effects o varying the slotlength and slot width on the S curve are shown. Figure(b)shows the effect o changing the length o slot 4(shown in

Figure). Te simulated S results or three different valueso the slot length are evident that changing the length oslot 4 (4= mm, 4 = mm, and 4 = mm) effectsthe extreme resonant bands by and large. Te effect on thelower resonant band is minor but the higher resonant band isalmost shifed completely, which is avorable in cases wherehigher resonant bands need a shif. On the other hand, themiddle resonant band has remained xed. In Figure(c), theS results show that changing the slot width also affects theS curve. It is clear that another way to shif the resonantbands could be just by varying the slot width. Te S resultsare presented or three different values o slot width, that is,= . mm, =.mm,and = . mm. Te resonant bandsat higher requencies shif their positions in the simulated Scurve and the lowest band remains unchanged.

Furthermore, to improve the efficiency and gain anotherenhancement we used was to insert the vacuum gaps insidethe dielectric material which we denote as substrate tem-pering. Te width and position o the vacuum gaps affectthe S curve which is shown in Figure (d). For threedifferent values o vacuum gap widths, that is, = mm, = .mm, and = mm, the S results show that


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S11(d

B)

Frequency (MHz)

With vacuum gapsWithout vacuum gaps

780 800 820 840 860 880 900

4

5

6

7

8

9

6.25 dB804 MHz

34 MHz

838 MHz 849 MHz

52 MHz 901 MHz

(a)

Gain(dB)

Frequency (MHz)

Simulated gain with vacuum gapsSimulated gain without vacuum gaps

Measured gain with vacuum gapsMeasured gain without vacuum gaps

0.81 0.84 0.87 0.90

0

2

4

2

6

dB3

(b)

F : (a) Bandwidth comparison or the simulated substrate tempering at a resonant requency band centered at MHz. (b) Gaincomparison with and without substrate tempering or the resonant requency band centered at MHz.

can offer a minimum SAR. In Figures (c) and (d), thegain at . GHz and . GHz is . dB and . dB,respectively. Te directivity o the proposed model at higherrequencies has a roughly omnidirectional shape which isessential or many handheld communication devices workingin this requency range.

In Figure, the magnitude o the E-eld at the corre-sponding resonant requencies is shown. From these gures,we can determine the path ollowed by the current orevery resonant requency band. Different paths ollowed bythe current on the radiating sheet are evident o the actthat inserting slots in the radiating patch provides multiplepaths or the current to ow and thereore gives rise tomultiple resonant requencies. Furthermore, these resonantrequencies and corresponding bandwidths can be enhancedwith shorting pins, parasitic patch, and other techniques.

.. Bandwidthsand Corresponding Gains. In this section, thebandwidths, resonant requencies, and gain or those corre-

sponding bands o requencies (st, nd, rd, andth resonantbands) are discussed. Te results show a wide bandwidth atthe resonant requencies. Tese are the optimized results oall the enhancements that we have applied to the proposedmodel which are discussed in Sectionin detail.

In Figures (a) and (b), a comparison o the bandwidthsand corresponding gains is presented. Te optimized resultor the proposed model at the st resonant band (lowestresonant band) shows a difference in the bandwidth andgain with and without tempering o the substrate. It is clearthat inserting the vacuum gaps in the substrate, in orderto exploit the impedance bandwidth , not only provides awide bandwidth but also helps to improve the gain. In our

proposed model, the simulated results show an increment inthe bandwidth rom MHz to MHz and gain is increasedand stabilized. One should choose the position and width orthe vacuum gap wisely (as already mentioned in Section.,position is important because it denes whether the resonantrequency o the cavity is increased or decreased).

Te gain is nearly at and stable or all these resonantband o requencies shown in Figure and is preerred bymost communications devices that work in this range orequencies.

4. Conclusion

In this paper, a new design or a low prole PIFA modelis presented. In general, applications may include com-munication devices that work or GSM /, UMS////, LE /, and ISM bands used or Bluetooth and WLAN. Te design is uniqueand simple. In contrast to a traditional PIFA model, thisdesign is covered with PEC boundaries rom all sides. Byintroducing a ew slots in the radiating patch, applying aparasitic patch, tempering the substrate, and using multipleshorting pins in the model, our resonant bands centered at MHz, MHz, MHz, and MHz have beenachieved with bandwidths o MHz, MHz, MHz,and MHz, respectively. Te gain or the correspondingresonant bands is relatively at. Multipleaspects o thisdesignwere studied which we have presented in this paper, and itis evident that this model has the ability to maintain its per-ormance even in adverse and unriendly environments. Teproposed design methodology can be useul in low prolemultiband resonant communication devices, in particular, inthe design o antennas or mobile handsets.


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0.0

1.5

3.0

4.5

1.70 1.72 1.74 1.76

Frequency (GHz)

Gain versus frequency

Gain(d

B)

(a)

0

3

3

6

1.95 2.00 2.05

Frequency (GHz)


Gain(d

B)

(b)

0

10

5

5

10

2.2 2.52.3 2.4 2.6

Frequency (GHz)


Gain(dB)

(c)

F : (a) Gain offered by the resonant band centered at . GHz. (b) Gain offered by the resonant band centered at . GHz. (c) Gainor the corresponding resonant band centered at . GHz.


Te authors declare that there is no conict o interestsregarding the publication o this paper.

Acknowledgment

Tis research was supported by the Research Fund BK pluso Kyungpook National University in .

References

[] H. . Chattha, Y. Huang, and Y. Lu, PIFA bandwidth enhance-ment by changing the widths o eed and shorting plates,IEEE

Antennas and Wireless Propagation Letters, vol. , pp. ,

.[] C. W. Chiu andF. L. Lin, Compact dual-bandPIFA with multi-

resonators,Electronics Letters, vol. ,no. ,pp., .

[] C. See, H. Hraga, and R. A. Hameed, A low-prole ultra-wideband modied planar inverted-F antenna,IEEE ransac-tions on Antennas and Propagation, vol. , no. , pp. ,.

[] N. Haider, D. Caratelli, and A. G. Yarovoy, Recent develop-ments in recongurable and multiband antenna technology,International Journal o Antennas and Propagation, vol. ,Article ID , pages, .

[] A. Cabedo, J. Anguera, C. Picher, M. Ribo, and C. Puente,Multiband handset antenna combining a PIFA, slots, and


38/132


ground plane modes, IEEE ransactions on Antennas andPropagation, vol. , no. , pp. , .

[] J. S. McLean, A re-examination o the undamental limits onthe radiation Q o electrically small antennas, IEEE ransac-tions on Antennas and Propagation, vol. , no. , pp. ,.

[] D. F. Sievenpiper, D. C. Dawson, M. M. Jacob et al., Experi-mental validation o perormance limits and design guidelinesor small antennas,IEEE ransactions on Antennas and Propa-gation, vol. , no. , pp. , .

[] M. Koubeissi, M. Mouhamadou, C. Decroze, D. Carsenat, and. Monediere, ribandcompactantennaor multistandardter-minals and users hand effect, International Journal o Antennasand Propagation, vol. , Article ID , pages, .

[] . ashi, M. S. Hasan, and H. Yu, Design and simulationo UHF RFID tag ant

antenna designing

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