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Ph.D. DISSERTATION
A STUDY ON MULTI-OCTAVE
GAN POWER AMPLIFIER USING
REACTIVELY MATCHED GAIN
CELL
리액티브 정합된 이득 셀을 사용한 다중 옥타브
GaN 전력증폭기에 관한 연구
BY
HONGJONG PARK
AUGUST 2018
SCHOOL OF ELECTRICAL ENGINEERING AND
COMPUTER SCIENCE COLLEGE OF ENGINEERING
SEOUL NATIONAL UNIVERSITY
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i
Abstract
In this thesis, a study on two-stage reactively matched gain cells
are proposed to implement a high-gain multi-octave distributed
power amplifier (DPA). The analytic analysis of proposed high-gain
distributed amplifier (DA) is presented and supported with simulation
and measurement results. Also, a shared bias network using simple
microstrip line is introduced. The bias network not only enables the
use of high-gain structures in DA configuration, especially in
monolithic microwave integrated circuit (MMIC) with compound
semiconductor process, but also has advantage on layout efficiency.
To further enhance the RF performances and circuit reliability, the
layout technique with reduced thermal coupling is applied. Finally, the
high power amplifier module implemented with four MMIC dies,
thanks to its high efficiency and low junction temperature.
The DA analysis starts off with lossy m-derived configuration in
artificial transmission lines, considering the effects of the gate
coupled series capacitor used in DPA. After that, reactively matched
cell, the proposed high-gain structure, is introduced and analyzed
using Thévenin equivalent. The gain of two DPAs, conventional
cascaded DPA and proposed reactively matched distributed amplifier
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(RMDA) is then compared to derive the design criteria of the
reactively matched cell. The simulated results are presented to show
the potential advantage on achieving high gain performance.
The biasing of transistors is one of the common difficulties in
MMIC design with compound semiconductor process. The standard
compound semiconductor process only consists two metal layers,
making the interconnection of the bias network to be difficult.
Therefore, the most of the gain enhancement techniques for DA are
implemented using CMOS process. The shared bias network is
proposed to overcome the layout limitation. Simple microstrip lines
are used for the proposed bias network, which behave similar to the
ideal bias network: short at dc, open at radio frequency (RF). Precise
analysis with active load modulation is done to design the shared bias
network. The shared bias network is verified with the RMDA
structure, and could be employed for other topologies.
The electro-thermal effect of GaN high power amplifier is also
studied. Power amplifiers (PAs) operating in a wide bandwidth have
limited efficiency and GaN power amplifiers dissipate large amount
of dc power. Therefore, the electro-thermal effect of a multi-octave
GaN power amplifier should be considered for circuit performance
and stability. Thermal coupling reducing approach and heat spreading
approach are presented and verified by measured results.
The implemented RMDA with the compact transistor layout has
been implemented in a small die size of 10.7 mm2 and shows output
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powers reaching 40.3-43.9 dBm, power added efficiencies (PAEs)
of 16–27% and small signal gains of 15.3–23.2 dB. The RMDA with
the reduced thermal coupling achieves 40.6–43.4 dBm with a peak
PAE of 29% in a slightly larger die size of 13.8 mm2.
Keyword: Broadband amplifier, distributed amplifier (DA), GaN
monolithic microwave integrated circuit (MMIC), multi-octave, high
power amplifier (HPA)
Student Number: 2012-20778
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iv
Contents
Abstract ....................................................................................... i
Contents ..................................................................................... iv
List of Figures ........................................................................... vi
List of Tables............................................................................ xii
Chapter 1 Introduction ............................................................. 1
1.1 Motivation .......................................................................... 1
1.2 Outline of This Thesis ....................................................... 7
Chapter 2 Multi-octive, High-Power, and High-Gain
Distributed Amplifier Structure Using Reactively Matched
Cells .................................................................................... 9
2.1 Introduction ....................................................................... 9
2.2 Previous Work and Proposed Structure for High-Gain
Distributed Power Amplifier ............................................. 12
2.3 Analysis of Gain of Proposed Reactively Matched
Distributed Amplifier ......................................................... 20
2.4 Design criteria and Detailed Design of Reactively Matched
Distributed Amplifier ......................................................... 37
2.5 Conclusion ....................................................................... 44
Chapter 3 Shared Bias Network for Proposed Reactively
Matched Distributed Amplifier Structure .......................... 45
3.1 Introduction ..................................................................... 45
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3.2 Shared Bias Network Using Simple Microstrip Lines ..... 48
3.3 Analysis of the proposed bias network ........................... 53
3.4 Multi-section analysis and detailed design ..................... 61
3.5 Conclusion ....................................................................... 65
Chapter 4 Chip Layout with Consideration of Thermal Coupling
.......................................................................................... 66
4.1 Introduction ..................................................................... 66
4.2 Design of Standard and Staggerd Layout Technique ...... 69
4.3 Measurements of the Fabricated MMICs ........................ 76
4.4 Conclusion ....................................................................... 92
Chapter 5 High Power Amplifier Module Combining Four RMDA
MMICs ............................................................................... 93
5.1 Introduction ..................................................................... 93
5.2 Design of Power Dividing and Combining Structure ....... 95
5.3 HPA Module Fabrication and Measurement .................. 103
5.4 Conclusion ..................................................................... 110
Chapter 6 Conclusions ......................................................... 111
Bibliography ........................................................................... 112
Abstract in Korean ................................................................ 118
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vi
List of Figures
Fig. 1.1. Schematic of multi-octave power amplifier structures,
(a) RMPA and (b) DA. ........................................................ 4
Fig. 1.2. Schematic of (a) a cascode DA and (b) an equivalent
FET expression of a cascode unit cell. ............................. 11
Fig. 1.3. Schematic of (a) a CSSDA and (b) an equivalent FET
expression of a CSSDA unit cell. ...................................... 13
Fig. 1.4. Schematic of a two-tier matrix DA. ......................... 17
Fig. 1.5. Schematic of a two-stage conventional cascaded
DPA. .................................................................................. 17
Fig. 1.6. Block diagram of the proposed distributed power
amplifier. ........................................................................... 19
Fig. 2.1. Schematic of a conventional distributed amplifier. .... 20
Fig. 2.2. Equivalent circuit of the input artificial transmission line.
.......................................................................................... 21
Fig. 2.3. Equivalent circuit of the transistor. ........................... 23
Fig. 2.4. A section of (a) constant-k line, (b) IATL, and (c)
OATL. ................................................................................ 27
Fig. 2.5. Distributed power amplifier using series capacitors. 29
Fig. 2.6. Block diagram of a cascade of two conventional
distributed amplifiers. ....................................................... 29
Fig. 2.7. Calculated power gain of distributed power amplifiers
with and without series capacitors. ................................... 36
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vii
Fig. 2.8. Thevenin equivalent of the interstage of the reactively
matched cell. ..................................................................... 36
Fig. 2.9. Calculated power gain of distributed power amplifiers
with series capacitors. ...................................................... 38
Fig. 2.10. Calculated design criteria of the reactively matched
cell. .................................................................................... 38
Fig. 2.11. Schematic of the interstage matching network. ...... 40
Fig. 2.12. Simulated matched impedance contour. .................. 41
Fig. 2.13. Voltage reflection coefficient and effective gain of the
reactively matched cell. .................................................... 41
Fig. 2.14. Calculated gain of the proposed distributed power
amplifier compared to that of the conventional cascaded
distributed power amplifier. .............................................. 43
Fig. 3.1. Example of the dc bias network problem in the proposed
RMDA. ............................................................................... 47
Fig. 3.2. Schematic of a self-biasing unit cell. ........................ 47
Fig. 3.3. Concept of active load-pull. ...................................... 49
Fig. 3.4. (a) Schematic and (b) equivalent circuit of the two-
section reactively matched distributed amplifier. ............. 49
Fig. 3.5. Plot of (3.8). .............................................................. 52
Fig. 3.6. Calculated added admittance of the simplified bias
network in the two-section reactively matched distributed
amplifier. ........................................................................... 57
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viii
Fig. 3.7. Calculated change of effective gain of the interstage of
the reactively matched cell caused by the simplified bias
network in the two-section reactively matched distributed
amplifier. ........................................................................... 60
Fig. 3.8. Detailed schematic of the proposed distributed power
amplifier. 200 Ω resistors are added to the gate bias
circuits of both drive and main stages to guarantee the
unconditional stability. ...................................................... 62
Fig. 3.9. Simulated results of the maximum magnitude value of
added admittance of the simplified bias network in the eight-
section reactively matched distributed amplifier. ............. 63
Fig. 3.10. Simulated added admittance from the bias network of
an eight-section reactively matched distributed
amplifier. ........................................................................... 64
Fig. 4.1. Measured junction temperature of a GaN HEMT device
according to dissipated power level, showing the self-
heating effect. ................................................................... 67
Fig. 4.2. Measured thermal coupling effect of a PA circuit. .... 67
Fig. 4.3. Layout of the proposed DPA MMIC with standard layout.
.......................................................................................... 70
Fig. 4.4. Layout of the proposed DPA MMIC with staggered
layout. ................................................................................ 71
Fig. 4.5. Conceptual diagram comparing the standard layout and
staggered layout in terms of thermal coupling. ................. 73
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ix
Fig. 4.6. The simulated temperature distribution of die surface
of standard layout (left) and the staggered layout
(right). ............................................................................... 74
Fig. 4.7. Photograph of the fabricated reactively matched
distributed amplifier with standard layout. ....................... 77
Fig. 4.8. Photograph of the fabricated reactively matched
distributed amplifier with staggered layout. ..................... 78
Fig. 4.9. Photograph of the test fixture to test the proposed
amplifiers. .......................................................................... 79
Fig. 4.10. Block diagram of the (a) test fixture and (b) heat sink
configuration for RF characterization the fabricated
MMICs. .............................................................................. 80
Fig. 4.11. The measured temperature distribution of die surface
of standard layout (left) and the staggered layout
(right). ............................................................................... 81
Fig. 4.12. Measured and simulated performances of small-signal
S-parameters of the proposed distributed power amplifier
with the standard layout. ................................................... 82
Fig. 4.13. Measured and simulated performances of output power
and efficiency of the proposed distributed power amplifier
with the standard layout. ................................................... 83
Fig. 4.14. Measured and simulated performances of small-signal
S-parameters of the proposed distributed power amplifier
with the staggered layout. ................................................. 84
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x
Fig. 4.15. Measured and simulated performances of output power
and efficiency of the proposed distributed power amplifier
with the staggered layout. ................................................. 85
Fig. 4.16. The simulated temperature distribution of die surface
of standard layout with CVD diamond thermal spreader (left)
and without CVD diamond thermal spreader (right). ........ 86
Fig. 4.17. Measured and simulated performances of small-signal
S-parameters of the proposed distributed power amplifier
with the CVD diamond thermal spreader. ......................... 87
Fig. 4.18. Measured and simulated performances of output power
and efficiency of the proposed distributed power amplifier
with the CVD diamond thermal spreader. ......................... 88
Fig. 5.1. Block diagram of the proposed high power amplifier
module. .............................................................................. 94
Fig. 5.2. The proposed microstrip-to-coaxial 1:2 divider and (b)
the conventional microstrip-to-coaxial divider. .............. 96
Fig. 5.3. Cross-sectional diagram of (a) the proposed
microstrip-to-coaxial 1:2 divider and (b) the conventional
microstrip-to-coaxial divider. ......................................... 97
Fig. 5.4. Simulated electric field of (a) the proposed microstrip-
to-coaxial 1:2 divider and (b) the conventional microstrip-
to-coaxial divider. ............................................................ 98
Fig. 5.5. Simulated S-parameter of (a) the proposed
microstrip-to-coaxial 1:2 divider and (b) the conventional
microstrip-to-coaxial divider. ......................................... 99
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xi
Fig. 5.6. Schematic of 50-Ω-to-100-Ω Wilkinson
combiner. ......................................................................... 101
Fig. 5.7. (a) Layout of the proposed Wilkinson combiner and (b)
simulated S-parameter of the proposed Wilkinson combiner.
........................................................................................ 102
Fig. 5.8. Sections of half module: (a) transition section, (b)
Wilkinson section, (c) MMIC section, (d) Wilkinson section
and (e) transition section. ............................................... 104
Fig. 5.9. (a) Photograph and (b) measured insertion loss of the
fabricated 1:2 microstrip-to-coaxial transition. ............ 105
Fig. 5.10. (a) Photograph and (b) measured insertion loss of the
fabricated total 1:4 power divider. .................................. 105
Fig. 5.11. Photograph of the fabricated HPA half-module. ... 106
Fig. 5.12. Photograph of the fabricated HPA module. ........... 107
Fig. 5.13. Photograph of the mersurement set-up of HPA
module. ............................................................................ 107
Fig. 5.14. Measured large-signal performances of the fabricated
HPA. ................................................................................ 108
Fig. 5.15. Measured large-signal performances of the fabricated
HPA in 18 GHz. ............................................................... 109
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List of Tables
TABLE 1.1. Si, GaAs, SiC, and GaN Material Properties .......... 2
TABLE 1.2. Parameters for calculating the equivalent
transconductance of the CSSDA unit cell ......................... 14
TABLE 2.1. Parameters values of transistors under class-AB
bias condition .................................................................... 33
TABLE 2.2. Designed values of components of interstage
matching network of reactively matched cell ................... 40
TABLE 4.1. Performance comparison table of GaN broadband
PAs .................................................................................... 91
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Chapter 1
Introduction
1.1. Motivation
From the beginning of the last century, electronic warfare (EW)
systems have contributed to protecting friendly forces and
weakening enemy weapons. The role of electronic warfare will
increase even more with the development of technology and the
change of battlefield environment. With the advent of network-
centric warfare (NCW), it is essential to secure the electromagnetic
spectrum for the acquirement of information assets and command and
control (C2). Also, cyber electronic warfare (CEW) technology is in
development to connect to a wireless network to perform operational
activities.
To meet the battlefield needs, the development of high-power
transmitter for the electronic attack is important. Conventional
traveling wave tube amplifiers (TWTAs) have many disadvantages:
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bulky and heavy, long heat-up and cool-down time, and short
lifetime. In addition, digital signal processing and wide bandwidth
operation cannot be incorporated with the TWTA. Therefore, the
solid-state power amplifiers (SSPAs) based on semiconductor
technology, which can overcome many drawbacks of TWTA, have
been actively developed to replace the TWTA for electronic warfare
applications.
Table 1.1 compares the physical properties of semiconductor
devices for fabricating SSPA [1]-[2]. Since GaN has wide energy
TABLE 1.1
Si, GaAs, SiC, and GaN Material Properties
Property Si GaAs SiC GaN
Bandgap (eV) 1.11 1.43 3.2 3.4
Breakdown
Field (MV/cm) 0.6 0.65 3.5 3.5
Electron Mobility
(Cm2/V-sec)
1350 6000 800 1000
Saturated velocity
(107cm/sec)
1 1 2 1.5
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bandgap, high breakdown field, high electron mobility and high
saturation velocity, GaN HEMT devices have high power density,
high operating voltage level and high-frequency characteristics.
Therefore, SSPAs based on GaN HEMT monolithic microwave
integrated circuit (MMIC) process are most suitable as a substitute
for TWTAs.
Design of a wideband power amplifier (PA) is a challenging task.
High power performance is associated with large sized transistors,
but large sized transistor has high Q-factor, resulting in narrow
bandwidth. In particular, EW system typically requires PA of more
than 10 watts of output power with linear power gain of 15 dB, in
multi-octave bandwidth. Efficiency is also a feature of importance,
at least more than 10% in terms of the power added efficiency (PAE)
is necessary. Therefore, every aspect of the performance of the PA
should be carefully analyzed along with achieving a multi-octave
bandwidth. Possible circuit topologies for wideband high power
amplifier (HPA) are reactively matched power amplifier (RMPA) and
distributed amplifier (DA), shown in Fig. 1.1.
In the RMPA, the strategy of lowering the Q-factor of each
transistor is performed. Lossy components and negative feedback is
typically employed. However, these approaches adds unnecessary
RF signal loss, which leads to reduced gain and efficiency [3]-[5].
In addition, the size of the RMPA is larger than DA due to the output
matching network and power combining circuit. One of the most
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(a)
(b)
Fig. 1.1. Schematic of multi-octave power amplifier structures, (a)
RMPA and (b) DA.
RFin
RFout
input matching
network
output matching
network
Lossy match
RFin
RFout
input artificial transmission line
output artificial transmission line
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5
recent work employed the non-Foster matching concept to RMPA
[6]-[7]. The authors called their topology as negative impedance
matched power amplifier (NMPA), and managed to reduce the Q-
factor without lossy elements. However, the instability of the non-
Foster circuit still limits RF performance of the PA.
DA, often referred to the traveling-wave amplifier, is most
suitable topology for any type of broadband amplifiers. The input and
output capacitances of the transistor are absorbed into the artificial
transmission lines (ATLs), thus bandwidth limitation come from
transistor Q-factor can be overcome. DA had not been considered as
a power amplifier for a long time, but non-uniform distributed power
amplifier (NDPA) theory demonstrated the possibility of the
distributed power amplifiers (DPAs) [8]. Nevertheless, DPA suffers
low gain and severe gain roll-off at high frequencies.
In this thesis, a topology for high-gain DPA is introduced. The
proposed structure is called reactively matched distributed amplifier
(RMDA), because the reactive matching concept is used to enhance
the gain of DPA. The practical layout difficulty of dc bias network
aroused in complicated circuit topology is also studied. The shared
bias network is proposed to deal with the layout problem. Also, the
staggered layout technique is introduced to reduce the thermal
coupling of the transistors, further improving RF performances. A
high power amplifier module combining four MMICs is implemented
for higher output power, thanks to the high efficiency and lower
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junction temperature of the fabricated MMICs.
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1.2. Outline of This Thesis
This thesis is composed of four sections. In the first section,
chapter 2, the reactive matching technique for DPA is presented. The
chapter starts off with the analysis of the limitation of the gain
performance in the conventional DPA. The reactively matching
technique is introduced, and the gain of the reactively matched cell is
analyzed to derive the design criteria. The detailed design of
reactively matched cell is presented at last, and the gain of RMDA is
compared to that of the conventional DPA.
In chapter 3, the shared bias network is introduced. The shared
bias network consists of simple microstrip lines. The principle of
operation is analyzed with active load-pull analysis. The added
admittances caused by adding bias network is calculated. To meet
the condition that the bias network should not affect the RF
performance, real and imaginary parts of the added admittance need
to be zero. The case of non-zero added admittance is also analyzed
to clearly show the effect of the shared bias network.
Chapter 4 focuses on the junction temperature and the RF
performance of the RMDA. The efficiency limitation of the broadband
PA results in large amount of power dissipated into heat. The junction
temperature should be considered not only for circuit reliability but
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also for RF performance. The staggered layout technique is proposed
to reduce the thermal coupling of each transistor in MMIC. Precisely
designed thermal carrier using CVD diamond material is applied to
MMIC with standard layout, and compared to the MMIC with
staggered layout.
Finally the thesis ends with conclusions in chapter 5 which
summarizes the RMDA design techniques.
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Chapter 2
Multi-Octave, High-power, and High-
Gain Distributed Amplifier Structure Using
Reactively Matched Cells
2.1. Introduction
Solid-state wideband HPAs are essential component for
broadband applications such as EW systems. DAs and RMPAs have
been mainly used as multi-octave PAs. In the RMPA, lossy
components are typically employed for a high return loss and low Q-
factor at the interstage, which leads to the decreased efficiency. In
addition, the size of the RMPA is large due to output matching
network and power combining circuit. Therefore, the HPAs designed
by the methodology of RMPA typically shows low power density [1],
[9]. The non-foster matching concept has also been applied to the
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RMPA, but the RF performances are still in limited level [6]-[7].
The DA is the most suitable circuit topology for a broadband
amplifier because the input and output capacitances of the transistor
are absorbed into the artificial transmission lines (ATLs). Therefore,
the bandwidth limitation from Bode-Fano criteria can be overcome
[10]-[11]. However, DA has two disadvantages as PA: low power
and low gain. Output power of conventionally-configured DA is
limited, because the output port of each transistor cells doesn't look
an optimum power load impedance (Ropt). To solve the output power
problem, the NDPA theory was presented [8]. By tapering the
characteristic impedances of each section's drain lines, each
transistor look load impedance close to Ropt, and the output power
could be increased with sacrificing a little amount of output return
loss. The NDPA theory had become a standard topology for ultra-
wideband DPAs.
On the other hand, there are many previous efforts to overcome
the low gain characteristic of DA, yet none of any were decisive. Low
gain characteristic of DA implies low overall gain and gain roll-off at
high frequencies. To solve the low gain problem of the DA structure,
various topologies have been proposed, such as the matrix DA [12]-
[13], the cascaded single stage DA (CSSDA) [14], and the cascode
DA [15]-[17]. However, it is challenging to design an HPA over 10
W with previously proposed structures at 6-18 GHz. Thus,
enhancing gain of DPA have been studied with interest by many
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11
researchers continuously until nowadays.
(a)
(b)
Fig. 1.2. Schematic of (a) a cascode DA and (b) an equivalent FET
expression of a cascode unit cell.
RFin
RFout
2×Vdd
Cds2
Cgs2
Cds1
Cgs1
gm2
gm1
Cds2
Cgs1
Gm,cascode
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12
2.2. Previous Work and Proposed Structure for High-
Gain Distributed Power Amplifier
Previous works for high-gain DPA can be classified as two types:
multi-stage DA approaches and unit cell modification approaches.
First of all, the previous unit cell modification approach mainly
includes cascode DA and CSSDA.
In the cascode DA, cascode cells replace the common-source
transistors in the unit cells (Fig. 1.2). Cascode cells are superior to
the common-source not only for the gain but also for the output
power. However, Cascode cells are difficult to use in GaN monolithic
microwave integrated circuits (MMICs) due to the high operating
voltage. Also, the DA using cascode cells has relatively small stability
margin and high process sensitivity, which leads to lower yields when
a large number of circuits are needed for a system such as EW
applications [15]-[17].
The latter, CSSDA, is a DA in which the common-source gain
cell of DA is replaced by a cascade of single-section DA [14].
CSSDA has higher gain to that of DA, since it utilized cascade
configuration so that the gain is multiplied. However, CSSDA suffers
large mismatches at interstage, thus gain should be reduced for
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13
(a)
(b)
Fig. 1.3. Schematic of (a) a CSSDA and (b) an equivalent FET
expression of a CSSDA unit cell.
RFin
RFout
Cds2
Cgs1
Gm,CSSDA
Zint
Cgs1
gm1
gm2
Cds2
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14
bandwidth with gain-bandwidth tradeoff. CSSDA also has the
disadvantage of gain that comes from interstage termination
resistances. More obviously, the series capacitor technique, which
will be explained next chapter, has to be applied to the main stage
TABLE 1.2
Parameters for calculating the equivalent transconductance of the
CSSDA unit cell
parameter description
gm1 transconductance of the first stage
transistor
gm2 transconductance of the second
stage transistor
Zint synthetic impedance of interstage
artificial transmission line
ω radial frequency
ωg1 gate radian cutoff frequency of the
first stage transistor
ωc radian cutoff frequency of the
artificial transmission lines
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15
transistor in CSSDA design for efficient bandwidth. The CSSDA gain
cell of Fig. 1.3(a) can be simplified into an equivalent field effective
transistor (FET) as Fig. 1.3(b) with effective transconductance
Gm,CSSDA. With the parameter given by Table 1.2, Gm,CSSDA can be
derived as follows:
1 2 2int 1
, 2 2
1
1exp ( )
2
2 1 1
m m gd
m CSSDA
cg
g g Z A A
G
(1.1)
Ad1 is drain attenuation per section in first stage and Ag2 is the
gate attenuation per section in second stage. The term of Ag2 is
affected by gate series capacitor of the main stage, and it dramatically
reduces the gain. Therefore, CSSDA exhibits only small amount of
gain improved, and does not overcome high-frequency gain roll-off.
The gain limitation of main stage series capacitor will be analytically
described later.
The multi-stage approaches are generally matrix DA and
conventional multi-stage DPA. The matrix DA is a simple and useful
technique for high gain DA. In the topology, FET configuration is
two-dimensional, looks analogous to matrix (Fig. 1.4). Thanks to
using interstage ATLs, the amplifier can be multi-stage without
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16
many modifications on design of DA [12]-[13]. However, the
biggest obstacle that prevents matrix DA for DPA is interstage ATLs.
The nodes of interstage ATLs which connects two transistors look
large capacitance. Especially for DPAs, The value of drain
capacitance and that of the gate capacitances are both large, so the
added amount become too large to be absorbed into ATL. Therefore,
matrix DA is not suitable for DPA topology.
The multi-stage DPA is a fairly simple way to enhance gain. By
cascading conventional DAs with multi-stage, the gain of total circuit
becomes multiplication of the gain of cascaded DAs (Fig. 1.5).
Typically for EW applications in 6-18 GHz frequency, the multi-
stage DPA is configured as two-stage, the first stage with a higher
gain and the last stage with a higher output power. Though this
approach still suffers limited gain and high frequency gain roll-off, it
is commonly used for GaN DPA MMICs. Every reported works for
6-18 GHz GaN DPA MMIC are designed by conventional cascaded
DA topology [18]-[20].
The proposed topology utilizes reactively matched gain cell to
enhance gain efficiently. The concept of interstage impedance
matching of RMPA is employed to unit cell design of DA. The
proposed design concept belongs to the unit cell modification
approach. Two-stage reactively matched cells between the input
artificial transmission line (IATL) and the output artificial
transmission line (OATL) substitute common source stages. A
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17
Fig. 1.4. Schematic of a two-tier matrix DA.
Fig. 1.5. Schematic of a two-stage conventional cascaded DPA.
RFin
RFout
Cds1
Cds1
Cds1
Cds1
Cgs2
Cgs2
Cgs2
Cgs2
RFin
RFout
-
18
simple block diagram of the proposed DPA is shown in Fig. 1.6.
The proposed DPA is designed using a commercial 0.25-μm
GaN HEMT foundry process by WIN Semiconductors Corporation.
The process provides two metal layers, a Si3N4 insulating film for
metal-insulator-metal capacitor, TaN thin film resistor, and air-
bridge for passive component design. The HEMT device shows a
cut-off frequency (fT) of 30.1 GHz, an fmax of 74.6 GHz, and a
transconductance of 227 mS/mm.
To achieve output power higher than 10 W in the 6-18 GHz
bands, the size of the main stage transistor is determined to be
6×125 μm, and the number of sections is determined to be eight.
The transistor with a size of 6×125 μm has a maximum small-
signal gain of 12 dB at 18 GHz and a maximum output power of 34
dBm with 30 V bias based on the load-pull analysis. To drive power
with high efficiency at 18 GHz, a 4×75 μm-sized transistor, which
has a maximum output power of 30 dBm with an input power of 22
dBm, is used as the drive stage. The interstage reactive matching
network is employed in each gain cell. The OATL is designed using
an NDPA theory to enhance the output power. The circuit RF
performance analysis starts off with schematic-based simulation,
and the final layout design is optimized through the electromagnetic
simulations using ADS Momentum.
-
19
Fig. 1.6. Block diagram of the proposed distributed power amplifier.
-
20
2.3. Analysis of Gain of Proposed Reactively Matched
Distributed Amplifier
The gain of the conventional DA and the two-stage cascaded
conventional DPA is calculated to compare to that of RMPA. To
analyze the gain of DA, the closed-form gain expression of a
conventional DA (Fig. 2.1) is calculated using the simplified circuit
as derived in previous works [16]-[17].
Fig. 2.1. Schematic of a conventional distributed amplifier.
RFin
RFoutZout
Zin
Ld/2 L
dL
d
Lg
Lg
Lg/2
Ld/2
Lg/2
Ld/2
Lg/2
Ld/2
Lg/2
1 2 3 n
Vg1
Vg2
Vg3
Vgn
Ig1
Ig2
Ig3
Ign
Id1
Id2
Id3
Idn
Iout
-
21
The analysis of the gain of DA is started off with calculating the
current and voltage of each node, as shown in Fig. 2.1 [21]-[22].
First of all, to calculate the gate voltage of k-th section Vgk, the IATL
is simplified as Fig. 2.2. First of all, the current of the gate of k-th
section Igk can be derived as
2 12 gk
ingkI I e
, (2.1)
where θg = Ag + jΦg is the propagation constant of the IATL. On
the other hand, using the image impedance equation of IATL, Vgk is
derived as
Fig. 2.2. Equivalent circuit of the input artificial transmission line.
2Ri
2Ri
2Ri
2Ri
2Ri
2Ri
Cgs
/2 Cgs
/2 Cgs
/2 Cgs
/2 Cgs
/2 Cgs
/2
Lgs
/2 Lgs
/2 Lgs
/2 Lgs
/2 Lgs
/2 Lgs
/2Ig1Vg1 Ig2Vg2 IgnVgn
Zin
Vin
Iin
-
22
2
1
g g
ggk gk gk
c
L CV I Z I
, (2.2)
where ω is radian frequency and ωc = 2πfc = 2/(LgCgs)1/2 =
2/(LdCds)1/2 is the radian cutoff frequency of the ATLs. Also, the
relation between input current Iin and the input voltage Vin can be
expressed as
2
1
1
inin in
gTg g c
VI V
ZL C
. (2.3)
From (2.1), (2.2), and (2.3), Vgk is expressed according to Vin,
2 12
21
gk
ingk
c
eV V
. (2.4)
To derive the drain current of k-th section, the voltage across
the k-th section transistor, Vck is calculated. In this analysis, the
transistor is assumed to be unilateral, because the gate-drain
-
23
capacitance Cgd can be absorbed into Cgs and Cds. Also, the instability
from Cgd is simplified equivalent circuit of the transistor is reduced
due to the gate/drain termination of the ATLs. Vck is derived as
1tan
2
1
11
gjgs
ck gk gkgsi
g
j C eV V V
R j C
, (2.5)
where ωg = 1/RiCgs is the gate radian cutoff frequency.
The current of the output terminal, Iout is summation of the
forward-direction drain current of the k-th transistor, derived as
Fig. 2.3. Equivalent circuit of the transistor.
-
24
1 2
1
d dn n k
out dkk
I I e
, (2.6)
where θd = Ad + jΦd is the propagation constant of the IATL. Using
(2.4), (2.5), and Ids equation in Fig. 2.3, (2.6) is developed to follows:
1
12
1
1tan
2
2 2 1
12
2 1 1
d d
g gd dgd
n n kjmout ck
k
jn j n km in
kg c
I g e e V e
g V e e e ee
. (2.7)
Φd and Φg are constrained to be equal to IATL and OATL to have
the same cutoff frequency and to have the highest Iout. Letting Φd =
Φg = Φ, (2.7) is further developed as
1 1tan
2
2 2 12 1 1
gg d dgd
A Aj nAj jn n k A Am inout
kg c
g V e e e e eI e
-
25
1tan
2
2 2
sinh2
1sinh2 1 1
2
gg d
jj jn g nd A Am in
gg c d
nA A
g V e e ee
A A
.
(2.8)
The input power and the power delivered to the load of DA are given
by
2 2
21re 1
22in in g
cinggT
V V LP
CZ
and (2.9)
2 2
2re 1
2 2out out d
cout dTd
I I LP Z
C
, (2.10)
respectively. Therefore, the voltage gain can be calculated as
20
22 2
sinh2
12 1 1 sinh
2
g dn A A
m gg od d
DA
c g gk k d
ng Z Z A A e
A
x x A A
, (2.11)
-
26
where xk = ω/ωc is the normalized frequency and Z0g and Z0d are the
characteristic impedance of IATL and OATL, respectively.
To quantify Ad and Ad, propagation function analysis of a
constant-k line is performed [23]. From a section of a constant-k
line shown in Fig. 2.11(a), the propagation function can be
determined from following relation
12
cosh 12
Z
Z , (2.12)
where θ is the propagation function of the line. θ can be expressed
as θ = A + jΦ, where A and Φ are respectively the attenuation
and phase shift per section. From (2.12), Φ can be canceled out, and
A is given as
2 2
1 1
2 2
2 2
re 1 im 12 2
1cosh sinh
Z Z
Z Z
A A
. (2.13)
when A
-
27
1
2
2
1
2
im 12
1 re 12
Z
ZA
Z
Z
. (2.14)
Evaluating (2.14) for the specific circuits shown in Fig. 2.4(b) and
Fig. 2.4(c), the closed-form expressions for attenuation on IATL and
OATL is obtained:
(a) (b) (c)
Fig. 2.4. A section of (a) constant-k line, (b) IATL, and (c) OATL.
Z1/2 Z
1/2
Z2
Lg/2 L
g/2
Ri
Cgs
Ld/2 L
d/2
Rds
Cds
-
28
2
221 1
c g kg
c g k
xA
x
and (2.15)
21
cd
d
k
Ax
. (2.16)
where ωd = 1/RdsCds is the drain radian cutoff frequency.
When using the DA as a PA, it is required to use large transistors
to obtain sufficient output power. However, the cut-off frequency of
the IATL is reduced with the increase of the transistor size, due to
the increase in the gate-source capacitance (Cgs). Therefore, to
ensure the multi-octave operation bandwidth, design techniques
such as series gate capacitor have been widely used, as shown in Fig.
2.5 [24]-[25]. However, the gain-bandwidth product remains
constant with the series capacitor technique, implying that there
should be significant gain degrade to cover a wide bandwidth.
The gain of two-stage cascaded DPA using series capacitors is
calculated to show the difference in gain between the conventional
cascaded DA and the RMDA. If a series capacitance of Cs = qCgs is
used, gate effective gate capacitance, the capacitance seen from the
input of the transistor, becomes
-
29
Fig. 2.5. Distributed power amplifier using series capacitors.
Fig. 2.6. Block diagram of a cascade of two conventional distributed
amplifiers.
RFin
RFoutC
sC
sC
sC
s
Cgs
Cgs
Cgs
Cgs
-
30
1gs gsq
C Cq
. (2.17)
As a result of decrease in the effective gate capacitance, the cut-
off frequency is increased by the factor of (1+q)/q. On the other hand,
the voltage across the gate capacitance is decreased by the factor of
q/(1+q) due to the voltage dividing,
1m mq
g gq
. (2.18)
Therefore, the formula for the DA gain using gate series capacitor is
20 0
22 2
sinh2
12 1 1 sinh
2
gdn A A
m gg d d
DA
c g gk k d
ng Z Z A A e
A
x x A A
, (2.19)
where
2 2c
g gs d dsL C L C
, (2.20)
1g
gsiRC
, (2.21)
-
31
2
221 1
c g kg
c g k
xA
x
, and (2.22)
kc
x
. (2.23)
The altered terms by the series capacitance are denoted with
prime symbols. In the lower frequency sub-band, where xk ≤ 0.4,
A'DA is degraded into [q/(1+q)]A, namely the low-frequency gain
reduction ratio. The gain of the cascade of two DAs, each using the
series capacitors, can be expressed as
1 22
1 21 2
2 22 2 2
1 2
1 21 2
1 21 2
4
exp 2
1 1 1
sinh 2 sinh 2
sinh 1 2 sinh 1 2
outin intm mstage
g gd d
c cg gk k k
g gd d
g gd d
g g Z Z ZA
n A A A A
x x x
n A A n A A
A A A A
.(2.24)
The gain of the two-stage conventional cascaded DPA is
evaluated with the selected transistors and circuit component values
-
32
from the available technology. The cascaded DA circuit is composed
of two eight-section DAs, DA1 and DA2, as shown in Fig. 2.6. To
determine the size of the transistors, we have set the bias of the
transistors to optimize the RF performance and then modified the
transistor size to present the required Cgs for DA operation. Under
class-AB bias chosen for PAE consideration, Cgs, Cds, Ri, and Rds
values of transistors with a size of 4×75 μm are 0.6 pF, 0.13 pF, 4
Ω, and 300 Ω, respectively. The parameter values for 6×125 μm
transistors are 1.6 pF, 0.3 pF, 4 Ω, and 120 Ω. Z0g and Z0d are set
to be 50 Ω. These transistor parameter values are presented in
Table 2.1. For a 50 Ω IATL with a 20 GHz cut-off frequency, series
capacitance of 0.68 pF and 0.40 pF should be used. Whereas the
bandwidth of DAs increases to 20 GHz with series capacitors, the
low-frequency gain reduction ratio of DA1 is 0.53 while that of DA2
is 0.20. Gain of DA1 and DA2 with and without using the series
capacitor technique is numerically evaluated in terms of frequency,
as shown in Fig. 2.7. In terms of power gain, the gain reduction can
be as much as 19 dB for the conventional two-stage cascade design.
The close-form equation of gain of RMDA is also derived to
compare the gain with the conventional cascaded DA [26]-[27].
RMDA can potentially achieve similar or higher gain than the
cascaded DA due to the high gain of the reactively matched cells that
replaces the common source transistors. The reactively matched cell
consists of two transistors and an interstage matching network as
-
33
TABLE 2.1
Parameters values of transistors under class-AB bias condition
parameter description Value
Cgs1 Gate-source capacitance of
a 4×75 μm transistor 0.6 pF
Cds1 Drain-source capacitance of
a 4×75 μm transistor 0.13 pF
Ri1 Gate-source capacitance of
a 4×75 μm transistor 4 Ω
Rds1 Input resistance of
a 4×75 μm transistor 300 Ω
Cgs2 Gate-source capacitance of
a 6×125 μm transistor 1.6 pF
Cds2 Drain-source capacitance of
a 6×125 μm transistor 0.3 pF
Ri2 Gate-source capacitance of
a 6×125 μm transistor 4 Ω
Rds2 Input resistance of
a 6×125 μm transistor 120 Ω
-
34
shown in Fig. 1.6. To analyze the reactively matched cell, Thevenin
equivalent circuit of the interstage of the reactively matched cell is
shown in Fig. 2.8. Assuming that the interstage matching network is
lossless, power delivered to the load can be expressed as
2 2
1
11
8L ds dsP I R
, whereas (2.25)
2
2
2
12
gk
Lin
VP
R
. (2.26)
From (2.25) and (2.26),
2
22 1 1
11
2 ingk ds dsV I R R . (2.27)
Therefore, the equivalent transconductance of the reactively
matched cell, namely Gm, defined by Ids2=GmVck1, can be found as
-
35
21 2 21
2
2
1
2 1
m m indsm
g
g g R RG
. (2.28)
Define Ais, the effective gain of the interstage matching network as
21isA . (2.28)
Finally, replacement of gm in (2.19) with Gm in (2.28) results in the
explicit formula of gain of RMDA with respect to Ais,
1 2 21
1 2
2 22 2 2
1 2
1 2
1 2
4
exp 2
1 1 1
sinh 2
sinh 1 2
outinm m indsRMDA
g d
c cg gk k k
g d
g d
g g Z Z R RA
n A A
x x x
n A A
A A
. (2.29)
-
36
Fig. 2.7. Calculated power gain of distributed power amplifiers with
and without series capacitors.
Fig. 2.8. Thevenin equivalent of the interstage of the reactively
matched cell.
Rds1
Ids1
Rds1
ap1
bp1
Cds1
interstage
matching
network
bp2
Vgk2 V
ck2
Rin2
Cgs2
-
37
2.4. Design criteria and Detailed Design of Reactively
Matched Distributed Amplifier
The ratio of ARMDA and A2-stage in (2.29) and (2.24) provides the
design criteria for the reactively matched cell to achieve the gain of
RMDA to be higher than that of the cascade of the conventional DAs.
The ratio ARMDA / A2-stage is given by
22
2 22 1
22
1 22
22
1 22
1 21 2
1 21 2
11
exp 2
sinh 21
sinh 1 21
sinh 1 2 sinh 1 2
sinh 2 sinh 2
kRMDA
stageg d
c g dg k
g dc g k
g gd d
g gd d
xA q
A q n A A
n A Ax
A Ax
A A A A
n A A n A A
isA
.(2.30)
In the lower frequency sub-band, where xk ≤ 0.4, xk2 is zero
and (2.30) is approximated as [(1+q2)/q2]Ais. In the higher
-
38
Fig. 2.9. Calculated power gain of distributed power amplifiers with
series capacitors.
Fig. 2.10. Calculated design criteria of the reactively matched cell.
-
39
frequency sub-band, (2.30) is a monotonically decreasing function.
Therefore, the condition of Ais ≥ [(1+q2)/q2] is the interstage
matching network design criteria for RMDA to achieve higher gain
than that of conventional cascaded DAs. Fig. 2.9 compares the
calculated gain of an ideal RMDA (without any interstage mismatch)
with the cascaded DA, using the parameters defined earlier.
To make the gain of the proposed DA larger than that of the
conventional cascaded DA, there are the minimum effective gain and
the maximum voltage reflection conditions, defined by ARMDA / A2-stage
≥ 1 in (2.30). The minimum effective gain and the maximum voltage
reflection for the reactively matched cell is calculated in Fig. 2.10. It
is observed that as long as the voltage reflection coefficient is less
than 0.85, the gain of the RMDA can be higher than the cascaded DA.
Following paragraphs show the detailed design of reactively matched
cell to meet the design criteria.
Unlike the case of a general RMPA, where four or more
transistors are connected in parallel, the reactively matched cell is
composed of one drive transistor and one main transistor. Thus, gate
port of the main transistor in reactively matched cell does not present
excessively small impedance. Therefore, it is possible to use reactive
lossless matching to cover a multi-octave bandwidth in RMDA.
The schematic of the reactively matched cell is shown in Fig.
2.11. The specific values of designed components are presented in
Table 2.2. With the chosen values, the mismatch was minimized in 7
-
40
Fig. 2.11. Schematic of the interstage matching network.
TABLE 2.2
Designed values of components of interstage matching network of
reactively matched cell
component Value
Ls 1.2 nH
Cblk1 8.3 pF
Cp 0.4 pF
Cblk2 8.3 pF
width of TLs 80 μm
length of TLs 200 μm
width of TLp 10 μm
length of TLp 280 μm
-
41
Fig. 2.12. Simulated matched impedance contour.
Fig. 2.13. Voltage reflection coefficient and effective gain of the
reactively matched cell.
-
42
GHz, 13 GHz, and 18 GHz for 6-18 GHz operation frequency band,
without any lossy matching technique or negative feedback. Za and
Zb, the impedance looked at both direction of node N in Fig. 2.11, is
simulated through ADS simulator with provided process design kit to
show the impedance matching at interstage of reactively matched cell.
Simulated Z*a and Zb are shown in Fig. 2.12. It can be observed that
the impedance matching is optimized in 13 GHz and 18GHz,
intentionally mismatching 7 GHz to have higher effective gain in
higher frequencies to deal with the gain roll-off. The voltage
reflection coefficient Γ and the effective gain Ais are also simulated
to compare to the design criteria, as shown in Fig. 2.13. It is evident
that proposed RMDA structure using the reactively matched cell
exhibits superior gain performance than the conventional cascaded
DA. The closed-form expression of the gain of RMDA (2.29) is
calculated using the simulated result of the reactively matched cell
(Fig. 2.14). Not only is the problem of the overall gain reduction
mitigated, but the gain roll-off at higher frequency edge is also
avoided due to the reactive matching technique.
-
43
Fig. 2.14. Calculated gain of the proposed distributed power amplifier
compared to that of the conventional cascaded distributed power
amplifier.
-
44
2.5. Conclusion
In this chapter, the concept of RMDA is proposed. Through
analytic analysis and numerical computation, the limitation of the
small-signal gain of the conventional cascaded DPA is presented.
The component that reduces the gain the most is specified, which is
the series capacitor of main stage transistor. To overcome the gain
limit, the reactively matched cell is proposed as a unit gain cell of the
distributed amplifier. Thanks to the reactively matched cell, IATL
composed of the main transistor can be removed, and the gain
reduction by the series capacitor of the main transistor is abolished.
The proposed reactively matched cell also has the benefit of having
two transistors only, thus interstage matching is done without any
lossy elements or negative feedback. Comparing the explicit formula
of conventional cascaded DPA and the proposed RMDA, the design
criteria of the RMDA is obtained. The detailed design of RMDA is
performed, and the simulated results clearly exhibits higher gain than
that of the conventional cascaded DPA, which implies that the
theoretical analysis is verified by the circuit simulation.
-
45
Chapter 3
Shared Bias Network for Proposed
Reactively Matched Distributed Amplifier
Structure
3.1. Introduction
In this chapter, the concept of shared bias network is proposed.
In the active circuit, every transistor needs dc bias for proper
operation. In the RF circuit, the bias circuit should be carefully
designed to feed dc current whilst decoupling RF signal. To achieve
RF decoupling, various techniques are proposed. Connecting a low
pass filter, RF bypass capacitor with a quarter-wave transmission
line, and RF choke inductor are common example of the biasing
circuits.
For the RFICs and MMICs, there are some limitations for biasing
-
46
networks. Biasing networks should be compact to save the size of
total circuit. Especially MMIC with compound semiconductor
technology, there are only two metallization layers. In fact, those two
metal layers are for air-bridge for RF signal: two metal layers should
be merged to feed high RF current in PA circuit. When designing a
circuit with complex topologies, it is almost impossible to feed dc bias
to "inside" transistor (see dotted red box shown in Fig. 3.1).
To overcome the biasing problem, a self-biasing scheme using
series feedback is proposed [28]-[29]. The self-biasing scheme
uses a series feedback resistor and a series feedback capacitor.
Voltage drop across the series feedback resistor produces gate-
source voltage. Series feedback capacitor provides short impedance
at RF signal, thus RF signal loss could be minimized. The schematic
of self-biasing is shown in Fig. 3.2. Although this technique is
beneficial for biasing the transistors, the voltage drop across the
resistor generates substantial dc power dissipation. Therefore, RF
performance degradation is unavoidable.
In this chapter, a compact shared DC bias network is proposed to
apply a single VDD bias to all transistors in the drive stage of RMDAs.
The proposed bias network is also applicable to any type of
distributed amplifiers or parallel power combining PAs.
-
47
Fig. 3.1. Example of the dc bias network problem in the proposed
RMDA.
Fig. 3.2. Schematic of a self-biasing unit cell.
gatedrain
-
48
3.2. Shared Bias Network Using Simple Microstrip
Lines
Given that the bias network should not affect the RF performance,
the design of the bias network had been focused on making RF open
circuit. However, in the presence of more than one RF signals, the
impedance is altered with the other RF signal. The active load-pull
concept explains the effect, as shown in Fig. 3.3. The impedance seen
at the junction node Z1 is expressed as
11
2
1I
Z RI
. (3.1)
Using this equation, the impedance in RF frequencies could be
maximized for bias network. In this thesis, a bias network consists of
simple microstrip lines are designed through active load-pull concept.
The two-section RMDA with the interstage drain bias line is
shown in Fig. 3.4. To understand the influence of the phase of the
bias line on the RF characteristics of the circuit, the case of a two-
section RMDA is briefly analyzed. Let the phase delay of a section of
-
49
Fig. 3.3. Concept of active load-pull.
(a) (b)
Fig. 3.4. (a) Schematic and (b) equivalent circuit of the two-section
reactively matched distributed amplifier.
Gen 1 Gen 2
R
I1
I2
V
Z1
IN
OUT
Qd1
Qd2
Qm1
Qm2
φ θ
X1
X2
I1
I2
I11
I12
Y1
Y2
YL
YL
θ
X1
X2
I1
I2
I11
I12
Y1
Y2
YL
YL
-
50
the bias line be θ, and the phase delay of a section of the IATL be
φ. In such case, I11 and I2 become
11 1jI kI e , and (3.2)
2 1jI I e , if (3.3)
1 2I I . (3.4)
Using the active load-pull equation in (3.1), Y2, the admittance
at node X2, can be expressed as
22
11 2
(1 )L LI
Y Y A jB YI I
, (3.5)
where
2
2
cos( )
1 cos( )
k kA
k k
and (3.6)
2
sin( )
1 cos( )
kB
k k
. (3.7)
To make Y2 to be identical to YL, which is in the case of the ideal
-
51
bias network, the conditions A = 0 and B = 0 should be satisfied.
Therefore, k = 0 and φ = θ could be a solution.
In terms of physical meaning, k means the ratio of amount of
leakage RF current to that of incident RF current. Therefore, the
value k cannot be zero, but has to be minimized. On the other hand,
φ is given as a phase delay of a section of IATL, θ is possibly
designed to be equal to φ.
With the condition of φ = θ, (3.7) is zero regardless of k. The
added susceptance value (imaginary of Y2−YL) is given as BYL, and
becomes zero with the condition. The analysis shows a design criteria
for the length of the microstrip line of the proposed bias network. On
the other hand, minimizing the added conductance value (real of
Y2−YL) is not that simple. With the phase condition of above analysis,
A becomes
2
21
k kA
k k
, (3.8)
which is plotted in Fig. 3.5. It is clear from Fig. 3.5 that k should
be minimized to make the added conductance lower. One of the
methods of reducing k is utilizing the current division. The
characteristic impedance of the microstrip line of the bias network
should be minimized. In addition, connecting the bias network to a
-
52
node that the impedance of load looked lowest is beneficial to
minimize k. The added conductance value at X2 is negative, implying
that the current is injected from the bias network. On the other hand,
the added conductance value at X1 is positive, meaning that the
current is leaked to the bias network.
In conclusion, the simple design criteria is acquired through
simple analysis. The design criteria is applicable to the practical
circuit design, but lacks detailed analysis and the effect of non-zero
added admittance. The precise non-zero added admittance analysis
and multiple load impedance effect will be discussed in next chapters.
Fig. 3.5. Plot of (3.8).
-
53
3.3. Analysis of the proposed bias network
Given that the bias network should not affect the RF performance,
the impedance and phase of each bias line should be designed to
minimize the added admittance from the bias network itself. The
design criteria derived in the previous chapter is further specified in
this chapter, with the quantitative analysis.
The schematic of two-section RMDA shown in Fig. 3.4(a) and
the equivalent circuit for nodes X1 and X2 shown in Fig. 3.4(b). Let
the phase delay of the bias line connecting the two sections be θ,
and the phase delay of a section of the IATL be φ. Unlike the
analysis in previous chapter, both load impedances are considered
and the ratio k is not used. Instead, the characteristic impedance of
the bias line be Z0, and the ratio between Z0 and ZL is defined as k =
Z0/ZL. Assuming no reverse wave at the bias line, Y1, the admittance
seen at node X1, is Y1 = YL+Y'L, where Y'L is
2
tan
tanL Lk j
Y Yk jk
, (3.9)
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54
the load admittance of second section looked at node X1. I12 is
given as
12 1jL
L L
YI I e
Y Y
, (3.10)
by current division. Developing (3.10) with (3.9), the equation of
I12,
12 1 2tan
2 1 tan
jk jI I ek j k
(3.11)
is attained. Y2, the admittance seen at node X2 is given by the
following due the the active load-pull effect:
22
12 2L
IY Y
I I
(3.12)
by the active load-pull.
I1 and I2 denote the output current of drive stage in first section
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55
and second section, respectively. In the high-efficiency PA design,
the drive stage is almost in saturated power level. Therefore, the
output power of each drive transistor in RMDA exhibits
approximately identical current. Thus, the assumption of |I1|=|I2|
could be validated. With the assumption, I2 is expressed by
2 1jI I e . (3.13)
The added admittance Y2-YL is derived from (3.12),
122
12 2L L
IY Y Y
I I
. (3.14)
In the previous chapter, the added admittance is separated into
the added conductance and the added susceptance. In this chapter,
(3.14) is analyzed to provide insight to the shared bias network
design.
Substituting (3.11) and (3.13) in (3.14), the closed-form
expression of added admittance is expressed as a function of k and
θ:
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56
2 L L
C jDY Y Y
A jB
, where (3.15)
cos tan sin 2A k k , (3.16)
2tan cos sin 1 tanB k k , (3.17)
cos tan sinC k , (3.18)
tan cos sinD k , and (3.19)
. (3.20)
The magnitude of (3.15) is given by
2 22
tan
cos sinL Lk
Y Y Y
, where (3.21)
2 4 2 25 2 2 tank k k , (3.22)
2 2 24 2 1 tank k , and (3.23)
2 22 ( 1)tank k . (3.24)
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57
The magnitude of added admittance, should be minimized for the
proposed bias network to behave as a dc bias network. The value of
φ in the practical RMDA design in previous chapter is 63°. (3.21)
is calculated as a function of θΔ and k in Fig. 3.6. In terms of k, the
larger the k, the smaller the added admittance. The result shows that
the characteristic impedance of the microstrip line should be as high
Fig. 3.6. Calculated added admittance of the simplified bias network
in the two-section reactively matched distributed amplifier.
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58
as possible, matching the analysis of previous chapter. However, the
minimum value for added admittance does not correspond to θΔ =
0 due to the transformed load impedance of adjacent section, Y'L.
Although the minimum added admittance appear at θΔ = 30°, the
condition of θΔ = 0 is more beneficial in terms of operation
bandwidth.
To understand the effect of the non-zero value of the added
admittance, the effective gain of the reactively matched cell, Ais, is
calculated with the presence of added admittance of the bias network.
When the added admittance is zero, the change in Ais becomes zero.
The assumption of the perfectly matched interstage was made for the
worst case, which is the case of the largest decrease in Ais. For
analytical calculation of Ais, |Y2+Y*L| = |Y2+YL| is assumed. The
assumption is valid when |Y2-YL| is small enough compared to
|Y2+YL|, as in our case of broadband reactive interstage matching.
The magnitude of voltage reflection coefficient of the interstage of
the reactively matched cell |Γ| is given as
2
2
L
L
Y Y
Y Y
. (3.26)
The calculated gain reduction with respect to θΔ and k is plotted
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59
as shown in Fig. 3.7. If the interstage of the reactively matched cell
is perfectly matched at a certain frequency, then Ais = 1 (0 dB). Let
the magnitude of the added admittance be 0.3 of the load admittance,
which is in the case of θΔ = 0 and k = 1.2, then the magnitude of
Fig. 3.7. Calculated change of effective gain of the interstage of the
reactively matched cell caused by the simplified bias network in the
two-section reactively matched distributed amplifier.
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60
the voltage reflection coefficient |Γ| is 0.195, and Ais is 0.962
(−0.17 dB). The gain drop due to the added admittance is minimal in
this case, which shows that the proposed bias scheme provides
leeway in the practical design.
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61
3.4. Multi-section analysis and detailed design
The analysis of two-section RMDA can be expanded to the
multi-section RMDA. The principle of superposition is applied to
estimate the added admittance value in the multi-section RMDA
shown in Fig. 3.8. Among nodes X1–8, the nodes X2–7 is the junction of
the bias lines upwards and downwards, the magnitude of the added
admittance is expected to cancel out. Therefore, it can be assumed
that the added admittance is nearly zero at the nodes X2–7, and the
maximum value of added admittance is present at node X1 or X8.
In the case of eight-section RMDA, the effect of the proposed
bias network is examined by the RF circuit simulation. The simulation
of eight-section RMDA is done by sweeping θΔ and Z0, and the
maximum magnitude of added admittance for the nodes X1 to X8 of
each case is shown in Fig. 3.9. The simulated data follow the general
trend of two-section RMDA in Fig. 3.6. However, there is some
discrepancy. This comes from the load admittance differences; the
load admittance is assumed to be constant in the two-section
calculation, but the simulated results account for the actual load
admittance in the eight-section RMDA (see Fig 2.12). However, one
can still draw the similar conclusion. The added admittance of less
than 15 mS has minimal impact on the effective gain for 6–18 GHz
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62
operation. Therefore, the phase error below 15° and the
characteristic impedance above 75 Ω allows insignificant gain
degradation.
In the fabricated circuit, the bias lines are designed to have the
same phase shift as the input transmission line, which is 63° at 18
GHz, to ensure the broadband operation. The characteristic
Fig. 3.8. Detailed schematic of the proposed distributed power
amplifier. 200 Ω resistors are added to the gate bias circuits of both
drive and main stages to guarantee the unconditional stability.
Qd8
OUT
IN
X1
X8
X7
X6
X5
X4
X3
X2
Qd1
Qd7
Qd6
Qd5
Qd4
Qd3
Qd2
Qm1
Qm8
Qm7
Qm6
Qm5
Qm4
Qm3
Qm2
Qd1–d7
Qd8
Qm1–m8
: 4×75 μm: 2×125 μm: 6×125 μm
VD1
VD2
λ/4
λ/4
Two-stage
reactively
matched
cells
Shared
bias
network
Odd-numberdinductors
Even-numberdinductors
VG1
VG2
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63
impedance is tapered from 105 Ω to 75 Ω, considering the current
handling capability of the metals used in the shared bias line. The
magnitude of added admittance is less than 13 mS as shown in Fig.
3.10. The added admittance is almost zero at the middle sections, as
expected from the principle of superposition. Thus, the fabricated
circuit is expected to show negligible performance degradation due
Fig. 3.9. Simulated results of the maximum magnitude value of added
admittance of the simplified bias network in the eight-section
reactively matched distributed amplifier.
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64
to bias sharing.
Fig. 3.10. Simulated added admittance from the bias network of an
eight-section reactively matched distributed amplifier.
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65
3.5. Conclusion
In this chapter, the shared bias network is proposed. To make
open circuit in RF domain, the proposed biasing scheme is analyzed
with the active load-pull theory. The design criteria for proposed
bias network is provided. Through quantitative analysis, the effect of
added admittance is calculated. Also, the change in the interstage
effective gain is calculated and simulated with respect to the added
admittance. Through the analysis, the possibility of shared bias
network is verified. The amount of added admittance and change in
the interstage effective gain is negligible for RF performance of total
DPA circuit.
The proposed bias network is applied to the RMDA in this thesis.
The shared bias network is also applicable to any other type of
circuits. For example, the biasing technique can be employed to
previously mentioned DA circuits, such as CSSDA. In addition, in the
case of the power combining circuit with finite phase delay, such as
balanced amplifier, the shared bias could be used for compact biasing
scheme. Therefore, the proposed bias technique is the possibly the
useful scheme for variety of MMICs in compound semiconductor
process.
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66
Chapter 4
Chip Layout with Consideration of Thermal
Coupling
4.1. Introduction
In the HPAs with a solid-state device, especially GaN PAs, the
junction temperature needs to be considered not only for circuit
reliability but also for RF performance such as power, gain, and PAE.
Also, common GaN PA produces large amount of RF output power
with significant amount of dc dissipation power. Due to the dissipated
power, GaN HEMT device heats itself while operating, as shown in
Fig. 4.1. It is widely known that the high junction temperature results
in the lower RF drain current. Therefore, the electro-thermal model
of the GaN HEMT is composed of a normal large signal model and a
thermal sub-circuit to predict the RF performance with respect to
junction temperature. In addition, to embrace the self-heating effect,
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67
Fig. 4.1. Measured junction temperature of a GaN HEMT device
according to dissipated power level, showing the self-heating effect.
Fig. 4.2. Measured thermal coupling effect of a PA circuit.
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68
the thermal resistance, defined as rise in junction temperature per
dissipated power, is included in thermal sub-circuit.
The thermal coupling effect is another major factor for increase
in the junction temperature in the MMIC as shown in Fig. 4.2. The
thermal coupling effect refers to the case where the heat generated
in each transistor is coupled to the adjacent transistors. With the
presence of the thermal coupling, the junction temperature of the PA
MMIC can be much higher than the single transistor cases.
To reduce the junction temperature in the bare-die circuit level,
Thermal carriers with various materials are proposed, but the
materials with higher thermal coefficient are costly. Self-heating can
be minimized by enhancing the efficiency, but in the case of the HPA
with multi-octave bandwidth, PAE is limited to about 30%. Therefore,
in this chapter, the layout technique with consideration of thermal
coupling is presented.
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69
4.2. Design of Standard and Staggered Layout
Technique
The layout of RMDA designed through chapter 2 and chapter 3
is performed. With consideration of thermal coupling, two types of
layout is proposed: the standard layout and the staggered layout. The
former focuses on the layout efficiency, and the latter emphasizes on
reducing thermal coupling. For a fair comparison, the RF performance
of two types are designed to be identical.
The layout of the standard layout is straightforward. First of all,
the layout of reactively matched cell is performed to fit between
IATL and OATL. The room for shared bias network should be left for
bias layout. Each reactively matched cell is in identical layout, except
the last section. In the last section, the size of the drive transistor is
altered to 2×125 μm, due to a finite amount of incident RF signal at
the shared bias. The layout with standard configuration is shown in
Fig. 4.3. Die size is 2.1 mm × 5.1 mm, the compact layout is possible
by the virtue of the shared bias network.
To reduce the effect of thermal coupling in the MMIC, a thermally
optimized transistor layout is also implemented. The layout with
staggered configuration is shown in Fig. 4.4. The reactively matched
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70
Fig. 4.3. Layout of the proposed DPA MMIC with standard layout.
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71
Fig. 4.4. Layout of the proposed DPA MMIC with staggered layout.
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72
cells are placed in a staggered configuration in Fig. 4.4 to increase
the physical distance between the adjacent main transistors. To place
the second-stage transistors far from the first-stage transistors,
the inductor of the reactively matched cell is replaced by a long high-
impedance microstrip line in the odd-numbered sections. On the
other hand, the even-numbered section places the second-stage
transistor closer to the first-stage transistor so that the second-
stage transistors can be staggered. The die size has been increased
from 10.7 mm2 to 13.8 mm2 (2.7 mm × 5.1 mm). As the main
transistors are separated, the thermal coupling effect can be
mitigated, and the junction temperature of the circuit can be reduced
toward the similar level of the single devices.
The conceptual diagram of comparing the standard layout and the
staggered layout is shown in Fig. 4.5. In the standard layout, the
minimum separation between the main-stage transistors is 431 μm
while it has been increased to 772 μm in the new layout. As shown
in Fig. 4.5, the equi-temperature contour encloses all of the main-
stage transistors in the standard layout. On the other hand, the
contour is separated in the staggered layout, implying that thermal
coupling is less significant in the case.
For the comprehensive electro-thermal design, the thermal
simulation has to be performed with circuit simulation, as in the case
of the EM simulation and schematic simulation. The thermal
simulation is performed with both layout, under the condition of DPA
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73
operation. The dissipated power is set to 60 W in accordance with
the circuit simulation results. The boundary conditions for thermal
Fig. 4.5. Conceptual diagram comparing the standard layout and
staggered layout in terms of thermal coupling.
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74
analysis are determined correspondingly to the test fixture. The
thermal simulation is performed assuming the PA die is attached to
5-mm-thick Au-plated Cu carrier with a high-thermal-conductive
adhesive. The ambient temperature is assumed 45 ℃ at the bottom
of the thermal carrier with convection.
From the simulated result shown in Fig. 4.6, it can be seen that
Fig. 4.6. The simulated temperature distribution of die surface of
standard layout (left) and the staggered layout (right).
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75
thermal coupling effect has been reduced using the proposed layout
technique. The equi-temperature contour of 100 ℃ in the standard
layout encompasses all the main-stage transistors, showing thermal
coupling. On the other hand, it is isolated to each device in the
staggered layout. The thermal resistance of the total circuit is
reduced from 2.8 ℃/W to 2.6 ℃/W, respectively. The simulated
maximum temperature is also reduced from 219 ℃ to 210 ℃.
Attained thermal resistance values of each transistor are
incorporated to the electro-thermal model of GaN HEMT, and RF
simulation is revised with the simulated thermal resistance. The
effect of the reduced thermal coupling can also be seen in the
improved RF performance, as shown in the following section.
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76
4.3. Measurements of the Fabricated MMICs
Two RMDAs were fabricated using a commercial 0.25-μm GaN
HEMT foundry process by WIN Semiconductors Corporation. The
first RMDA is fabricated with a standard layout and the thermal
coupling reduced layout technique is applied to the second RMDA.
The photograph of the fabricated MMICs are shown in Fig. 4.7 and
Fig. 4.8. The same electrical design has been applied to both RMDAs
to compare the effect of thermal management. Fabricated MMICs are
attached to an Au-plated Cu carrier with a high-thermal-conductive
adhesive, as shown in Fig. 4.9. The adhesive is EK2000 by Epoxy
Technology Inc. with a thermal conductivity of 35.5 W/mK. The test
fixture is composed of a 5-mm-thick Au-plated Cu carrier and a
heat sink with thermal grease, as shown in Fig. 4.10. The RF tests
are performed under continuous-wave conditions with a drain bias
voltage of 30 V.
On top of RF performance, the thermal profile of the RMDA with
the staggered layout is measured. To estimate the junction
temperature drop by the proposed layout technique, IR thermal
microscope image was taken to compare the thermal profiles of the
conventional layout and the staggered layout. Thermal measurements
were performed by setting the temperature of the bottom of the test
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77
Fig. 4.7. Photograph of the fabricated reactively matched distributed
amplifier with standard layout.
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78
Fig. 4.8. Photograph of the fabricated reactively matched distributed
amplifier with staggered layout.
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79
fixture at 45 ℃ and the dissipated power level at 60 W, in
accordance with the thermal simulation in previous chapter. The
measured results are shown in Fig. 4.11. The staggered layout shows
reduced thermal coupling and lower junction temperature compared
Fig. 4.9. Photograph of the test fixture to test the proposed
amplifiers.
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80
(a)
(b)
Fig. 4.10. Block diagram of the (a) test fixture and (b) heat sink
configuration for RF characterization the fabricated MMICs.
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81
to the conventional layout. It is worth noting that the measured
thermal profile is in close agreement with the simulated results in Fig.
4.6.
Fig. 4.11. The measured temperature distribution of die surface of
standard layout (left) and the staggered layout (right).
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82
The measured small-signal S-parameter results of the standard
RMDA are compared with the simulated S-parameter results in Fig.
4.12. From 6 to 18 GHz, a small-signal gain of 15.3-23.2 dB is
achieved while S11 and S22 are lower than -10 dB. The average
value of the measured small-signal gain is 20.1 dB, which is among
Fig. 4.12. Measured and simulated performances of small-signal S-
parameters of the proposed distributed power amplifier with the
standard layout.
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83
the highest gain reported from > 10W DAs. Output power and PAE
are shown in Fig. 4.13. The measurements were done by sweeping
the input power from 27 to 30 dBm at each frequency to achieve the
required output power and PAE. The output power of 40.3-43.9 dBm
and PAE of 15.5-26.6% are achieved with a small die size of 10.7
mm2. The average output power is 42.0 dBm, and the average PAE
is 21.0%.
Fig. 4.13. Measured and simulated performances of output power and
efficiency of the proposed distributed power amplifier with the
standard layout.
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84
The measured RF performances of the RMDA with thermal
coupling-reduced layout are shown in Fig. 4.14 and Fig. 4.15. The
measured data with the staggered layout demonstrate further
improvement in RF performances. The proposed RMDA exhibits a
slightly higher small-signal gain of 16.7-24.1 dB from 6-18 GHz,
with an average gain of 21.7 dB. The PAE performance is also
Fig. 4.14. Measured and simulated performances of small-signal S-
parameters of the proposed distributed power amplifier with the
staggered layout.
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85
improved, reaching 15.0-29.4%. The simulation is performed using
the device model provided by the foundry, which does not include the
effect of thermal coupling. The average PAE in 6-18 GHz
frequencies is 22.6%, which is enhanced by 1.6% due to the reduced