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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 22, NO. 1, JANUARY 2007 3

Estimation and Measurement of JunctionTemperatures in a Three-Level

Voltage Source ConverterThomas Brckner, Member, IEEE, and Steffen Bernet, Member, IEEE

AbstractThe design of a power converter must guarantee thatthe operating junction temperatures of all devices do not exceedtheir limits under all specified operating conditions. Usually, thisis ensured by a simulative or analytical junction temperature esti-mation based on simple electrical and thermal models and semi-conductor datasheet values. This paper discusses the difficultiesand quantifies the limitations of this approach on the example ofa three-level neutral point clamped voltage source converter (NPCVSC) with insulated gate bipolar transistors. The calculations arecompared to the results of direct junction temperature measure-ments with an infrared camera. The paper also provides the ex-perimental proof for the unequal loss and junction temperaturedistribution in the three-level NPC VSC.

Index TermsInsulated gate bipolar transistors (IGBT), Junc-tion temperature measurement, neutral point clamped voltagesource converter (NPC VSC), thermal modeling.

I. INTRODUCTION

FOR the design of a power converter the maximum electricalratings and the thermal limitations of the semiconductor

devices play an equally important role. It must be guaranteedthat the operating junction temperatures of all devices donot exceed their limits under all specified operating conditions.Therefore, the calculation of is an integral part of the con-verter design procedure. Nowadays, some semiconductor man-ufacturers provide online (internet) software tools for this pur-pose [1], [2]. Another recent development is the real-time calcu-lation of the operating junction temperatures to prevent thermaloverload[3],[4].

Not only in these two cases but also in the standard powerconverter design, the calculation is commonly based ondatasheet values for both, the electrical behavior (conduc-tion and switching losses) and the thermal characteristicsof the semiconductors. The level of the electrical modelingvaries from linear to third-order approximations. The semi-conductors thermal characteristics are typically described byone-dimensional (1-D) component models[5]. Whilethe electrical data can be verified by the means of uncompli-cated measurements, thermal device models are more difficult

Manuscript received November 3, 2005; revised March 29, 2006. Recom-mended for publication by Associate Editor J. A. Ferreira.

T. Brckner is with the Converteam GmbH, Berlin 12277 Germany (e-mail:[email protected]).

Steffen Bernet is with the Institute of Energy and Automation Tech-nology, Technical University of Berlin, Berlin 10587 Germany (e-mail:[email protected]).

Digital Object Identifier 10.1109/TPEL.2006.886651

to validate. Their parameters are obtained either bycomplex physics-based finite element method (FEM) devicemodels or by the measurement of via temperature-dependentelectrical semiconductor characteristics in special test circuits[6][8].

Though the calculation based on these simple electricaland thermal models is widely used, nearly no data are availablequantifying its accuracy and its limitations en bloc. Attempts

to experimentally verify the calculated junction temperaturesin a real converter under practical operating conditions are notknown to the authors.

Therefore, the common method for junction temperature cal-culation is analyzed in this paper. The approach is applied to athree-level neutral pointclamped voltage source converter (NPCVSC) with insulated gate bipolar transistors (IGBTs) and theestimated temperatures are compared to junction temperaturesdirectly measured in this converter. Thus, the difficulties, theaccuracy, and the limitations of the junction temperature esti-mation method as a whole can be evaluated.

A schematic of the three-level NPC VSC is shown inFig. 1.

Thetopologyhas been chosenas an example,because itsthermaldesign is more complicated than that of two-level converters.The NPC VSC displays an inherently uneven loss distributionamongst its semiconductor switches, dependent on the specificoperating points. The most unbalanced distribution can befound at the boundaries of the four-quadrant operating area, i.e.,at maximum 1.155 and minimum modulation depth

0 and power factors 1 (inverter operation) and1(rectifieroperation).Ineachofthesecasesonegroupof

devices experiences both, significant switching and conductionlosses, while all other devices are substantially less stressed[9]. The paper provides the experimental proof for the loss and

junction temperature distribution in the NPC VSC. Moreover,

the junction temperature measurements described in this paperare also used in[10]to validate the loss-balancing principle ofthe NPC VSC featuring active NPC switches (ANPC VSC).

In detail, a low-voltage (400 V, 10 kVA) test setup was con-structed whose function and design is described inSection IIofthe paper. The electrical and thermal modeling of the laboratorysetup is explained inSection III. The difficult task of the junc-tion temperature measurements was performed with the aid ofan infrared camera system on open IGBT modules as describedin Section IV. Following, the results of calculation and measure-ment are compared and evaluated in Section V. Special attentionis given to the accuracy and the problems associated with both

junction temperature measurement and prediction (Section VI).0885-8993/$20.00 2006 IEEE

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4 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 22, NO. 1, JANUARY 2007

Fig. 1. Three-level NPC VSC.

Fig. 2. Schematic of testsetup with two phase legsusedas

-bridgeconverter.

II. LOW-VOLTAGELABORATORYSETUP

To investigate the junction temperature distribution of the

three-level NPC VSC, a low-voltage test bench was set up. Itis configured as three-phase acdcac converter with 12-pulsediode-bridge rectifier and three-level inverter. A schematic ofthe inverter is shown inFig. 2.The basic converter data and rat-

ings are summarized in Table I. One phase leg of the inverter

is assembled of specially prepared open IGBT modules to en-

able the infrared junction temperature measurements. The open

modules are directly taken out of the production process prior to

the encapsulation and the filling with silicon gel. A second phaseleg is equipped with high-frequency Pearson current transducers

to measure the IGBT switching transients.

In low-voltage IGBT converters ( 200 V 690 V) the

devices are usually installed on a common heatsink or a highernumber of devices, e.g., a complete three-phase two-level con-

verter, is integrated into one power module. In this situation,

the common base temperature for the calculation of is the

heatsink or base plate temperature. The devices are thermally

strongly coupled. In contrast, in high-power medium-voltage

converters discrete devices are installed on separate heatsinks.

The devices are thermally coupled by the cooling water or am-

bient air only. Hence, the differences between the device junc-

tion temperatures can be larger than in a low-voltage converter.

An unequal loss distribution amongst the devices also yields an

unequal junction temperature distribution.

Since the three-level NPC VSC is typically applied in

high-power medium-voltage applications and its thermal char-acteristics are more critical in this situation, several measures

TABLE I

RATINGS OFLABORATORYTESTSETUP

are used to emulate the medium-voltage converters thermalbehavior in the low-voltage setup. Single IGBT modules are

not available for the low power rating of the test bench. There-

fore, each phase leg is assembled of three half-bridge modules

as shown in Fig. 2, viz. T T T T , and T T .For all experiments described in this paper, the IGBTs T

and T are not active and are permanently turned off by a

gate-emitter short.1 Their anti-parallel diodes D and D act

as NPC diodes. The arrangement of the modules guarantees

that none of them is stressed with more than 2 between

its terminals.

To thermally decouple the half-bridge modules from each

other, every module is installed on its own heatsink. Further-

more, it is desired to achieve a good thermal decoupling of the

two IGBTs in one module. For this purpose bipartite aluminum

blocks are inserted between the modules and the heatsinks.

Fig. 3 shows the mechanical layout. The aluminum blocksintroduce an additional thermal resistance between the junction

and thefirst point of a low-resistive common coupling, i.e., theheatsink. The coupling through the module base plate can not

be avoided. The temperature in the middle of each heatsink is

measured by a thermocouple.

To enable a good cooling and highest compactness (neces-

sary for the infrared measurements), the three heatsinks are in-

tegrated into one mechanical unit as depicted in Fig. 4. They

are mounted in series and are thermally insulated. HS-23 de-

notes the heatsink for the module with the IGBTs T and T ,

HS-15 for T and T , etc. The interaction between the three

heatsinks through the cooling air stream (heat convection) and

heat radiation remains.

For the thermal investigations, two phases of the test bench

were operated as four-quadrant -bridge converter with a

single-phase inductive load as indicated in Fig. 2. The load

resistance is the self resistance of the inductor. The phase

under test, i.e., the open phase, is voltage controlled by apure sine-triangle modulation with a given modulation depth

and fundamental frequency . The second phase

is used to generate the necessary countervoltage impressing a

sinusoidal current with the desired amplitude and phase

shift into phase one. The required modulation depth for

phase two and the phase shift of its fundamental voltage with

1The active NPC switches T and T are utilized in the experimentsfor the ANPC VSC described in [10].

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Fig. 5. Measured turnoff transient of T ( 2 325 V, 40 A,

15

, time 250 ns/div, voltage 100 V/div, current 20 A/div).

Fig. 6. Measured turnoff transient of T (conditions asFig. 5).

causes higher turn-off and smaller turn-on losses for T com-

pared to T .

The differences between the short and the long commuta-

tion were evaluated by switching loss measurements for both,

the recommended and the increased gate resistance. As an ex-

ample, the current and voltage waveforms of the aforementioned

turn-off transients of T and T with the gate resistance

15 are shown inFigs. 5and6, respectively. From the magni-tude of the voltage peak, the stray inductance in the commuta-

tion loops can be estimated to 180 nH for the short commuta-

tion circuit (turn-off of T ) and 360 nH for the long commu-

tation circuit (turn-off of T ). Measured switching losses for

15 and 56 are given inTable III. The in-

creased gate resistance roughly doubles the switching losses of

the IGBTs, while the diode recovery losseschange only slightly.

At the same time, the loss differences between the short and long

commutation are lessened. Nevertheless, they are considered in

the model.

It is found that the turn-on and turn-off losses

of the IGBTs display an almost linear dependence on the load

current. The recovery losses of the diodesare small com-pared to the IGBT switching losses. Therefore, the IGBT and

TABLE III

MEASUREDSWITCHINGLOSSES

Fig. 7. Measured IGBT and diode forward characteristics and approximations.

diode switching losses can be approximated with good preci-

sion by

(3)

where stands for either , or , or . The voltage

denotes the prospective commutation voltage of the de-

vices in the NPC VSC and is the voltage at which the

device loss measurements were performed at. is afittingconstant. Since the loss measurements were carried out exactly

at the level of the commutation voltage 2, both

and cancel from the equation.

Simple approximations for the IGBT and diode on-state pa-

rameters (threshold voltage and differential resistance) are given

in the datasheet. A series of on-state measurements shows that

these approximations deliver insufficient results for the rela-tively low currents ( 0 30 A) of interest for the low-

voltage test setup. Therefore, the measured IGBT and diode con-duction losses are approximated as

(4)

The measurements and their approximations are shown in Fig. 7.

(4)contains a logarithmic function of the on-state current mod-

eling the behavior of a pn junction and a resistive share for

the drift region of the semiconductor. Though the parameters

, and representa physical meaning for the char-acteristic of a diode (temperature voltage, saturation current, and

ohmic resistance of drift region, respectively), here they are not

more thanfitting constants. They were obtained by the meansof a standard curvefitting program.

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BRCKNER AND BERNET: ESTIMATION AND MEASUREMENT OF JUNCTION TEMPERATURES 7

Fig. 8. Thermal model of one half-bridge module with heatsink.

The blocking losses of the IGBTs and diodes as well as the

diode turn-on losses are negligibly small and, thus, not consid-

ered in the model.

B. Thermal Modeling

The thermal model for one half-bridge module with heatsink

is depicted inFig. 8. The purpose of the model is the calcula-

tion of the average junction temperatures. Thermal capacitances

are therefore not included. Compared to the usually applied

standard model the introduction of the aluminum blocks (see

Fig. 3) requires some refinements. Due to the increased resis-tance case-to-heatsink the coupling via the base plate must not

be disregarded. The network is extended by the transition resis-

tance between the two switches on the base-plate level (case-to-

case). In continuation of this concept also the heatsink is mod-

eled in two parts which are linked by a heatsink-to-heatsink re-

sistance. According to its actual position the thermocouple to

measure the mean heatsink temperature is placed in the middle

of this link. The two resistances case-to-heatsink include the

transition base plate-to-aluminum block, the resistance of the

block itself, and the transition aluminum block-to-heatsink.

The values of the thermal resistances are given inTable IV.

The case-to-case, heatsink-to-heatsink, and aluminum block re-

sistances are calculated from the specific thermal conductivityof the material and the geometrical data

(5)

and (6)

(7)

where istheaverage distancebetweenthesilicondies in the

module, and are the cross-sectional areas of the copper

base plate and the heatsink along the mirror axis, and ,

and are the height, length, and width of the aluminumblocks. 401 W/(m K) and 209 W/(m K) denote

TABLE IV

PARAMETERS OFTHERMALMODEL

the specific thermal conductivity of copper and aluminum, re-spectively.

The resistances for junction-to-case (IGBT and diode) are

taken from the datasheet. For the transitions between base plate,

aluminum block, and heatsink, thermal contact grease is applied

and the case-to-heatsink value from the datasheet is used.

The interaction of the three heatsinks via the cooling air is

considered by temperature-controlled temperature sources. The

dominant process is the mutual heating by convection through

the forced air stream. Assume the lowest heatsink HS-23 (see

Fig. 4) to be at a high temperature level. The ambient air passing

through HS-23 is heated and the cooling air temperature for the

heatsinks HS-15 and HS-46 is increased. Therefore, a fraction of

the heatsink temperature of HS-23 must be added to the ambient

temperature for the latter heatsinks

(8)

In equivalence to (8) linear functions are formulated for all

mutual dependencies. The appropriate gains of the temperature

sources and the thermal resistances heatsink-to-air

were obtained experimentally. The IGBTs of one module at a

time were connected to a dc-current source to produce definedconstant losses . The temperatures of the three

heatsinks are measured and their ratios are evaluated

and

(9)

(10)

is the temperature of the directly heated sink

and is the increased temperature of the consid-

ered neighboring heatsink. The results of these measurements

are given inTable IVas well.

As expected the coupling occurs predominantly in the direc-

tion of the airflow, but also in the opposite direction some in-fluence can be seen. Beside convection neighboring heatsinkscan affect each other also by radiation in both directions. More-

over, it is interesting to note that the investigation yields dif-ferent values for though all heatsinks are identical. The

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Fig. 9. (a) Photograph of phase leg with open modules. (b) Infrared picture at operating point

0.85

1. (c) Infrared picture at operating point

0.15 0

1; common conditions: ambient temperature

25 C,

650 V,

30 A,

8 kHz sine-triangle modulation.

first heatsink in the air stream is the best cooled; the last one theworst cooled. This phenomenon is independent of the coupling

described above and can be explained by increasing air turbu-

lences.

The presented model allows the estimation based on the

ambient air temperature (to be manually measured at the air

inlet) or based on the measured heatsink temperatures .

To eliminate the potential error of the heatsink data, the heatsink

temperatures are taken as the basis throughout this work.

IV. MEASUREMENT OFJUNCTIONTEMPERATURES

An overview of methods to determine semiconductor oper-

ating junction temperatures is given in[11]and [12]. The po-

tential approaches include the indirect measurement via temper-

ature-dependent electrical parameters and the direct measure-

ment with contact or contact-less sensors.

Though the indirect measurement is often used for the valida-

tion of thermal device models[7],[8], the approach is not suited

for the application under standard converter operating condi-

tions. Complicated changes in the control would be required to

include measurement cycles into the modulation scheme of a

real converter. Moreover, the tiny parameter variations, e.g., ofthe on-state voltage drop in the range of a few millivolts per

Kelvin, can not be resolved in the electromagnetic environment

of an operated pulse-width modulation (PWM) converter. A di-

rect measurement method is necessary that works independently

of the tested converter. The infrared temperature measurement

fulfils this requirement. It provides a strict separation of the con-verter and the measurement system, a high measurement speed,

and a high precision. Therefore, the authors decided for this

method.

The junction temperature measurements were performed

with an AGEMA Thermovision 900 professional infrared

imaging system [13]. The scanner camera with HgCdTe-de-

tector operates at a frame rate of 15 Hz with a resolution of136 272 pixels. A photograph of theopenphase leg as seen

by the camera and two infrared images at different operating

points of the converter are shown inFig. 9. The entire phase leg

with six IGBT and six diode chips is recorded in one camera

view. The three IGBT modules and the individual chips can be

clearly recognized inFig. 9(b)and (c).The chip areas are de-

fined in the processing software and the measured temperaturesare averaged over the individual chip areas.

A number of ambient measurement conditions have to be ob-

served. The sensor and the lens temperature are measured by the

camera and are automatically taken into consideration for thecamera-internal data processing. The ambient air temperature,

the humidity, the distance object-to-camera, and the emissivity

of the measured object must be entered into a protocol. Since

the emissivity of the silicon dies is unknown and for metallic

surfaces is typically dependent on the surface angle, a coating

of the IGBT modules is inevitable. It is realized with a thin layer

of black paint with a specified constant emissivity of 0.95.Since the camera operates in the spectral range of 8 12 m

(so-called atmospheric window with a transmission coefficientof the ambient air of 1), the recombination radiation of

the silicon semiconductors at a wave length of 1.1 m

isfiltered out. Note, that the recombination radiation does notdepend on the junction temperature but on the charge carrierdensity.

Fig. 10 shows a macroscopic infrared image of one IGBT

chip. It can be seen that the bond wires conceal parts of the

chip surface and, thus, their temperature contributes to the mea-

sured average chip temperature. To quantify the effect, a series

of macroscopic IGBT and diode measurements for various con-

ditions was carried out. It shows that the temperature of central

bond wires under stress is a few Kelvin higher than that of the

chips. However, the measured average temperature across the

chip surface (including the bond wires) is slightly lower than

the spot temperature in the center of the chip. The reason there-

fore is the lower temperature at the edges which are also partlycovered by passivation silicon gel. Considering both effects, a

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Fig. 10. Infrared image of a single IGBT chip under operation, average tem-perature 53.2 C, spot temperature 54 C.

Fig. 11. Measured junction temperature ripple versus time.

measurement error for the average temperature of the chip sur-

face of K remains.

The correct time averaging of the recorded data requires fur-

ther attention. Although the camera frequency is lower than the

desired output frequency of the converter, the beat frequency

effect can be used to measure the -time characteristic. Thefundamental frequency of the converter is set to 50.25 Hz.

With a camera frequency of 15 Hz the fundamental period (

19.9 ms) is sampled 20 times within 4 s if every third recorded

camera sample is used ( 200 ms). The result is depicted in

Fig. 11. The junction temperature ripple with can be clearly

recognized. It amounts to 2 K in the examined example.

As to how much this measured ripple equals the true ripple of the

junction temperature can not be evaluated. Though the silicon

die and the applied coating are very thin ( 220 m 30 m,

respectively), the tiny thermal capacitances of the chip and the

coating damp the ripple at the chip surface.

Moreover, the question arises as to how exact the measured

average temperature at the surface equals the average tempera-ture at the junction. Due to the heat radiation of the chip surface

a temperature drop across the coating is expected. The radiation

intensity is described by the StefanBoltzmann Law

(11)

with 5.67 10 W cm K . For a single IGBT chip

at 75 C ( 1 cm ) the total radiation amounts to 78.8 mW,

which is less than 0.2% of its total dissipated power. Hence, it

can be assumed that the temperature drop across the coating is

negligible small and the measured average temperature at the

surface equals the real average junction temperature.

V. RESULTS

The NPC VSC is investigated at the four operating points

specifiedin Table II. The measured temperatures (averaged overthe chip areas and the fundamental period) are compared to the

calculated temperatures inFig. 12. It is found that the estimated

values meet the measured values with tolerances of

1 5 K. In the average, the estimated temperatures of the

IGBTs and diodes are 0.6 and 1.7 K higher than the measured

IGBT and diode temperatures, respectively. This tendency indi-

cates a small systematic error. The maximum relative deviation

of the estimated to the measured values with respect to the am-

bient temperature amounts to 16.7%.

The comparison ofFig. 12(a)(d) also shows that in everyoperating point other groups of devices are stressed. The mea-

surements prove the fundamental behavior of the NPC VSC

as analyzed in[9] but also reveal some characteristics specificto the investigated low-voltage test bench. For inverter opera-

tion at large modulation depth [Fig. 12(a)] the outer switches

T and T are the most stressed devices, whereas at small

modulation depth [Fig. 12(c)] the NPC diodes D and Dexperience the highest temperatures. This result matches with

the general theoretical analysis. However, both operating points

for rectifier operation display a temperature distribution wherethe inner switches T /T and diodes D /D reach by far

the highest junction temperatures. At large modulation depth

[Fig. 12(b)] the outer inverse diodes D and D experience

both conduction and recovery losses, while the inner switches

are only stressed with significant switching and the inner diodeswith significant conduction losses. Nevertheless, the inner de-vices face the highest temperatures because they are joint in

one half-bridge module and their entire power losses have to

be carried off by only one heatsink. The situation for rectifieroperation at a small modulation depth [Fig. 12(d)], once again,matches with the general theoretical analysis. Here, the inner

switches are the most stressed devices.

Further discussion and comparison of the NPC VSCs lossand junction temperature distribution is given in[9]and[10].

VI. EVALUATION OF ERRORS

The good match of the measurement results with the esti-

mated temperatures must not lead to rash conclusions about the

accuracy of the calculation. A detailed analysis of errors is

given inTable V. The included estimates are based on semicon-

ductor datasheets, the discussion with manufacturers, and owncalculations.

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Fig. 12. Comparison of estimated andmeasured junction temperaturesat four operating points. (a) Case A

0.85

1.(b)Case B

0.70

0 1. (c) Case C 0.15 1. (d) Case 0.15 0 1. ( 25 C, 650 V, 30 A, 8 kHz, sine-triangle modulation).

The summation of all partial errors yields a total maximum

deviation of the estimated junction temperature from its truevalue of 8.25 9.29 K. This equals almost 30% of the mea-

sured junction temperature with respect to the ambient air tem-

perature. In this worst case scenario all measurement and mod-

eling tolerances affect the total error in the same direction. This

is possible, but rather unlikely. If the individual errors of the

loss approximations, thermal resistances, etc. are assumed to be

independent some errors can compensate each other. If the er-

rors are further assumed to be normal distributed, according to

the Gaussian law of error propagation the maximum total error

stays within 11% with a probability of 68%. This number cor-

responds to the practically achieved results inSection V.

The electrical and the thermal model contribute about equalshares to the total error. The electrical model is based on mea-

sured conduction and switching losses. Three major sources of

errors can be identified, viz. the measurements of the losses themselves; the missing reflection of the temperature dependence; the approximation of the losses.Though the loss measurements were performed with the

highest possible precision, the temperature dependence could

not be quantified. All measurements were carried out in therange of 25 45 C, whereas the operating junction tem-

peratures in the converter vary from 25 70 C. Thus, an

error of 10% for the effectof onthelosses mustbe assumed.

Also the linear switching loss approximations do not enable aprecision higher then 10%. If both, the loss approximations

are improved (error 5%) and the temperature dependence is

included (error 0%), the total maximum error of the estimatedjunction temperature can be reduced to 18 22%.

Whereas theelectrical model canbe improved with acceptable

efforts, there are no simple means to verify the parameters of

the thermal model. The thermal resistance for junction-to-case

given in the datasheet includes a safety margin for the aging

of the modules due to thermal cycling. Moreover, the whole

structure of the model bears the essential weakness of a radically

simplified geometry. E.g., the coupling of IGBT and diodethrough the substrate layer is not considered. The value

for the transition from the module base plate to the heatsink is

not more than a rough estimate. Here, the heat flow is affected

by a number of aspects as theflatness of the base plate as wellas the thickness and migration of the thermal contact grease.Since in the special situation of the test setup there are two

such transitions (from and to the inserted aluminum block)

this estimate clearly becomes the weak point of the thermal

model. Finally, the heat convection from the heatsink to the

ambient air is strongly dependent on the heatsink temperature,

the ambient temperature, air turbulences, etc. The datasheet

values of the heatsink do not describe the dependence on these

parameters in detail. Therefore, the error of the heatsink data is

eliminated as the measured heatsink temperature (accuracy

0.5 k) is taken as a basis for the calculation.

If the aluminum block and one of the transitions with thermal

contact grease are removed from the calculation, the totalmaximum error reduces to 20 24%. Considering a standard

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TABLE V

ERRORANALYSIS FORJUNCTIONTEMPERATURE ESTIMATION (EXAMPLE

CALCULATION FORT ATOPERATINGCONDITIONS OFCASEA )

mechanical design, i.e., without aluminum block, together

with an enhanced electrical model (see above) an accuracy of14 18% with respect to the ambient temperature can be

achieved. However, this precision can only be reached with

loss measurements for every individual device. A survey of

semiconductor datasheets shows that the typical manufacturing

tolerances yield a variation of the conduction and switching

losses in the range of 20 25%. Thus, a estimation based

on datasheet values can not deliver an accuracy higher than

about 25 30%.

As a final point, the accuracy of the junction temperature mea-surement is assessed. It is mainly determined by

the specified tolerance of the infrared camera ( 1 K);

the effect of the bond wires ( 1 K); the definition of the chip areas ( 1 K).The effect of the bond wires has been discussed in Section IV.

Further, it is estimated that the definition of the chip areas acrosswhich the temperatures are averaged causes another error of

1 K. Deviations from the assumed emissivity of the

chip surface can be disregarded due to the applied coating. Also

the temperature difference of the chip surface to the junction can

be neglected (see Section IV). Considering the three above men-

tioned factors a total measurement accuracy for absolute junc-

tion temperatures of 3 K can be concluded. The rel-

ative precision for the comparison of the junction temperatures

at different operating points against each other is significantly

higher since the definition of the chip areas remains unchangedand the specified resolution of the camera is 0.08 K.

VII. CONCLUSION

The authors investigate the junction temperature distribution

of a three-level NPC VSC with IGBTs by the standard cal-

culation method and by direct infrared measurements. Both ap-

proaches are described in detail. The unequal distribution of

the NPC VSC is experimentally proven.

However, close inspection yields that the analytical or sim-

ulative junction temperature estimation with simple electrical

and thermal ( component) models based on datasheet values

cannot deliver results with tolerances less than 25 30%. If

the datasheet values are complemented by precise loss mea-

surements in terms of current, voltage, and junction temper-

ature, an accuracy higher than 20% can be achieved. Since

there is currently no simple alternative this common estima-

tion method will be continuously used for the thermal design

of power converters. Due to its simplicity the discussed models

are also useful for the real-time calculation of junction temper-

atures[3], [4],[9]. Nevertheless, the users should be aware of

the tolerances and limitations.

ACKNOWLEDGMENT

The authors wish to thank Dr. U. Scheuermann and Dr. U.

Nicolai, SEMIKRON, for granting the open IGBT modules and

for valuable advice and discussion, and Dr. R. Schacht, Fraun-

hofer IZM, for providing the infrared camera system.

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Thomas Brckner (S99M06) received theDipl.-Ing. and Dr.-Ing. degrees in electrical engi-neering from the Dresden University of Technology,Dresden, Germany, in 1999 and 2005, respectively.

In 1996 and 1997, he was a Guest Student at theVirginia Power Electronics Center, Virginia Poly-technic Institute and State University, Blacksburg.From 1998 to 2005, he worked on several projects

for ABB in Germany and Switzerland as wellthe Technical Universities in Dresden and Berlin,Germany. In 2002, he was a Visiting Researcher at

Monash University, Clayton, Australia. Since 2006, he has been with CON-VERTEAM (former Alstom Power Conversion), Berlin. His research interests

include topologies, devices, and controls for high-power conversion.

Steffen Bernet (M97) received the M.S. degreefrom Dresden University of Technology, Dresden,Germany, in 1990and the Ph.D. degreefrom IlmenauUniversity of Technology, Ilmenau, Germany, in1995, both in electrical engineering.

During 1995 and 1996, he was a Postdoctoral Re-searcher in the Electrical and Computer EngineeringDepartment, University of Wisconsin, Madison. In

1996, he joined ABB Corporate Research, Heidel-berg, Germany, where he led the Electrical DriveSystems Group. From 1999 to 2000, he was respon-

sible for ABB research worldwide in the areas Power Electronics Systems,Drives,andElectric Machines.In 2001, he joined the Berlin University ofTechnology, Berlin, Germany, as a Professor of power electronics.

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