工學碩士學位論文ael.chungbuk.ac.kr/ael-results/theses/... · 2015-07-07 ·...

工

學

碩

士

學

位

論

文

소

형

화

된

R

F

I

D

리

더

기

안

테

나

CHINZORIG

2010年

2月

工學碩士學位論文

소형화된 RFID 리더기 안테나

A Miniaturized RFID Reader Antenna

忠北大學校大學院

바이오정보기술학과

친저릭 바토치르 (Chinzorig Bat-Ochir)

2010 年 2 月

工學碩士學位論文

소형화된 RFID 리더가 안테나


指導敎授安炳哲

바이오정보기술학과

친저릭 바토치르 (Chinzorig Bat-Ochir)

이 論文을 工學碩士學位論文으로 提出함.

2010 年 2月

本論文을 친저릭 바토치르의 工學碩士學位論文으로 認定함.

審査委員長 안 재 형 ㊞

審査委員 김 경 석 ㊞

審査委員 안 병 철 ㊞

忠北大學校大學院

2010 年 2月

-i-

CONTENTS

Abstract ······························································································································ ii

List of Figures ················································································································· iii

List of Tables ·················································································································· vi

I. Introduction ······························································································ 1

II. Antenna Design ··················································································· 3

2.1 Design Requirements ··············································································· 3

2.2 Literature Review ······················································································ 4

2.3 Proposed Antenna Structure ··································································· 8

2.4 Feed Network ························································································· 10

2.5 Radiating Elements ················································································ 23

2.6 Whole Antenna Structure ········································································ 34

III. Antenna Fabrication and Measurements ·········································· 41

3.1 Antenna Fabrication on PCB ································································· 41

3.2 Antenna Tuning ······················································································· 42

3.3 Antenna Measurements ··········································································· 44

IV. Conclusions ························································································· 48

References ·································································································· 49

Acknowledgements ···················································································· 50

-ii-


Chinzorig Bat-Ochir

Department of Bio & Information Technology

Graduate School, Chungbuk National University

Cheongju, Korea

Supervised by Professor Ahn, Bierng-Chaerl

Abstract

In this thesis, a miniaturized circularly-polarized RFID reader antenna operating at

912MHz is studied. For use in RFID readers, the antenna must be small and of light

weight while maintaining good performances. The weight of the existing ceramic patch

antenna is over 70g causing a major inconvenience to users of the portable RFID

reader. The purpose of this thesis is to develop a light-weight antenna with a

performance comparable to that of the existing ceramic patch antenna. The antenna

developed in this thesis is printed on FR-4 substrates with a total volume of 60×60×14

mm3. The antenna consists of a feed network and four radiating elements. A quadrature

hybrid coupler and two power dividers are used to realize the feed network. Four

horizontal monopole radiating elements are fed with a successive 90° phase difference

for circular polarization. The monopole is of an inverted-F type with short-circuited

impedance tuning stub. The monopole length is reduced by meander-line technique. In

order to reduce the feed network size, meander lines and a dielectric cover are

employed. The designed antenna is fabricated and experimentally tuned to obtain an

optimum performance at 912MHz. Measurements of the fabricated antenna show a gain

of 3.2dBic, an axial ratio of 0.6dB and a reflection coefficient of -16dB at 912MHz.

-iii-

List of Figures

Fig. 2.1 A typical portable RFID reader and its antenna ·········································· 3

Fig. 2.2 Geometry of a compact printed square QSA [1] ········································· 4

Fig. 2.3 Measured performance of QSA. (a) Return loss, (b) peak gain, and

(c) axial ratio [1] ······································································································ 5

Fig. 2.4 Measured radiation pattern of the single QSA at 910MHz [1] ·················· 6

Fig. 2.5 Arrow-shaped ring slot antenna for (a) linear and (b) circular

polarizations [2] ········································································································ 6

Fig. 2.6 Current distribution on the arrow-shaped ring slot antenna.

(a) Linear polarization and (b) circular polarization [2] ····································· 7

Fig. 2.7 Simulated and measured (a) reflection coefficient, (b) radiation pattern,

and (c) axial ratio of the arrow-shaped ring slot antenna [2] ··························· 8

Fig. 2.8 Proposed antenna structure ··············································································· 9

Fig. 2.9 Geometry of a quadrature coupler [3] ·························································· 10

Fig. 2.10 A simple quadrature coupler structure ························································ 11

Fig. 2.11 Scattering parameters of a simple quadrature coupler ······························ 12

Fig. 2.12 Phase of the transmission coefficient of a simple quadrature coupler · 12

Fig. 2.13 Phase difference of a simple quadrature coupler ······································ 13

Fig. 2.14 Miniaturized quadrature coupler ··································································· 13

Fig. 2.15 Cross section of the miniaturized quadrature coupler circuit.

(a) Board construction and (b) two boards bonded together ·························· 14

Fig. 2.16 50-ohm bended line of quarter-wavelength ·············································· 15

Fig. 2.17 Reflection coefficient of 50-ohm bended line ········································· 15

Fig. 2.18 Transmission coefficient of 50-ohm bended line. (a) Magnitude and

(b) phase ·················································································································· 16

Fig. 2.19 35-ohm bended line of quarter wavelength ··············································· 16

Fig. 2.20 Reflection coefficient of 35-ohm bended line ··········································· 16

Fig. 2.21 Transmission coefficient of 35-ohm bended line. (a) Magnitude and

(b) phase ·················································································································· 17

Fig. 2.22 Reflection coefficient of the miniaturized quadrature coupler ················· 17

-iv-

Fig. 2.23 Transmission coefficient of the miniaturized quadrature coupler ············ 18

Fig. 2.24 Transmission phase of the miniaturized quadrature coupler ···················· 18

Fig. 2.25 Transmission phase difference of the miniaturized quadrature coupler ·· 19

Fig. 2.26 Antenna feed network. (a) Circuit layout, (b) two boards unbonded,

and (c) two boards bonded together ···································································· 20

Fig. 2.27 Reflection coefficient of the feed network ················································ 21

Fig. 2.28 Transmission coefficient of the feed network ············································ 21

Fig. 2.29 Transmission phase of the feed network ···················································· 22

Fig. 2.30 Transmission phase deviation of the feed network ··································· 22

Fig. 2.31 Single radiating element ··············································································· 23

Fig. 2.32 Reflection coefficient of a single element versus the antenna length ···· 24

Fig. 2.33 Impedance matching of a single element by the shorting stub ·············· 24

Fig. 2.34 3D gain patterns of a single radiating element. (a) Total gain,

(b) gain of theta component, and (c) gain of phi component ························· 25

Fig. 2.35 2D total gain patterns of a single radiating element. (a) On zx-plane,

(b) on xy-plane, and (c)on yz-plane ····································································· 26

Fig. 2.36 Array of four radiating elements ································································· 27

Fig. 2.37 Change in the resonance frequency of a single element due to

mutual coupling ······································································································· 28

Fig. 2.38 Mutual coupling between antenna elements ··············································· 28

Fig. 2.39 3D gain patterns of the four-element array. (a) Total gain, (b) gain of

theta component, and (c) gain of phi component ·············································· 29

Fig. 2.40 2D total gain patterns of the four-element array. (a) On zx-plane,

(b) on yz-plane, and (c) on xy-plane ··································································· 30

Fig. 2.41 3D polarization patterns of the four-element array. (a) Axial ratio,

(b) right-hand circular polarization, and (c) left-hand circular polarization ···· 32

Fig. 2.42 2D axial ratio patterns. (a) On zx-plane and (b) on yz-plane ················ 33

Fig. 2.43 Final four-element antenna structure ··························································· 34

Fig. 2.44 Reflection coefficient of the final antenna ················································· 35

Fig. 2.45 Gain versus frequency of the final antenna ·············································· 35

-v-

Fig. 2.46 Axial ratio versus frequency of the final antenna ·································· 35

Fig. 2.47 3D gain patterns of the final antenna. (a) Total gain, (b) gain of

theta component, and (c) gain of phi component ·············································· 36

Fig. 2.48 2D total gain patterns of the final antenna. (a) On zx-plane,

(b) on yz-plane and (c) on xy-plane ···································································· 37

Fig. 2.49 3D polarization patterns of the final antenna. (a) Axial ratio, (b) right-

hand circular polarization, and (c) left-hand circular polarization ···················· 38

Fig. 2.50 2D axial ratio patterns of the final antenna. (a) On zx-plane and

(b) on yz- plane ······································································································ 39

Fig. 3.1 Photographs of the fabricated antenna components. (a) Feed network,

(b) radiating elements, and (c) tuning elements ················································· 40

Fig. 3.2 Photograph of the fabricated antenna ··························································· 41

Fig. 3.3 Antenna tuning by adjusting the length of radiating elements ················· 42

Fig. 3.4 Improvement in reflection coefficient by adjusting the length of radiating

element ····················································································································· 42

Fig. 3.5 Impedance tuning by modifying tuning strip width ··································· 43

Fig. 3.6 Reflection coefficient of the fabricated antenna ·········································· 43

Fig. 3.7 Gain versus frequency of the fabricated antenna ········································ 44

Fig. 3.8 Comparison of gain of the fabricated antenna with those of existing

antennas ···················································································································· 45

Fig. 3.9 Photographs of existing RFID reader antenna. (a) One by Actenna Co.

(size: 60x60x15mm), and (b) ceramic patch antenna by MAC Technologies

(size: 62x62x7mm) ·································································································· 45

Fig. 3.10 Measured gain patterns of the fabricated antenna. (a) On zx-plane,

(b) on yz-plane, and (c) on xy-plane ··································································· 46

Fig 3.11 Axial ratio versus frequency of the fabricated antenna ···························· 47

-vi-

List of Tables

Table 2.1 Design goal of the proposed RFID reader antenna ··································· 3

Table 2.2 Parameters of microstrip lines used in the miniaturized quadrature

coupler ························································································································ 14

Table 3.1 Performance of the fabricated antenna ······················································· 47

-1-

Ⅰ.Introduction

In the RFID system, the role of antenna is very important because the distance of

recognition is mainly determined by the antenna performance. Specially, the reader

antenna should have circularly polarization characteristics because the tag antenna can be

arbitrarily positioned. Most RFID reader applications require a circularly polarized

antenna of sufficient bandwidth and gain. A suitable antenna for portable terminals

should be of low cost, low profile, light weight and especially of small size.

Previous works have concentrated on several designs for RFID reader applications.

For microstrip antennas, some techniques, such as making slots or using high dielectric

constant substrates, can be used to reduce the antenna size. More than 50% size

reduction can be achieved, but it still cannot satisfy the requirement of light weight and

small size. The existing commercial antenna for the portable RFID reader application is

a ceramic patch antenna with a dimension of 62×62×7mm3, and a gain of 4.4dBic, but

it weighs over 70g causing a major inconvenience to users of the RFID reader.

In this thesis, a miniaturized light-weight circularly-polarized RFID reader antenna

operating at 912MHz is developed for use in portable RFID readers. The design goals

are as follow; volume within 62×62×15mm3, axial ratio less than 3dB, gain greater than

3dBic, and reflection coefficient less than -15dB. At 912MHz, the wavelength is 329mm

so that the required antenna volume is 0.19×0.19×0.05 wavelength.

In order to meet design goals, numerous antenna types are investigated. With the

given size requirement, most antenna types do not satisfy the gain and circular

polarization requirements. A final antenna type of choice is an array of four inverted-F

meander line monopoles fed with a successive 90° phase shift. Among many possible

schemes of feeding four radiating elements, a quadrature hybrid coupler combined with

two power dividers is selected.

First a single radiating element is designed using the widely-used commercial

electromagnetic software Microwave StudioTM (MWS) by CST. The antenna height, the

tuning stub and the meander line monopole radiator are designed. Secondly an array of

four radiating elements is analyzed by individually feeding the element with an idealized

source. The resonant frequency shift is investigated and the length of the radiator is

-2-

adjusted to compensate for mutual coupling between radiating elements. Thirdly the

antenna feed network is designed. In order to reduce the feed network size, microstrip

lines covered with a dielectric layer is employed. Lengths of transmission lines in each

arm of the quadrature hybrid are reduced by meander-line techniques. At two outputs of

the quadrature hybrid, a power divider is connected. In order to realize a successive

phase shift of 90°, the length of each arm of the power divider is adjusted. In the final

step, four radiating elements are fed by the feed network and the entire antenna

structure is simulated and optimized using MWS. The designed antenna has a volume of

60×60×14mm3 and a theoretical gain of 2.4dBic.

The designed antenna is fabricated using the common FR-4 substrate. The fabricated

antenna is experimentally tuned for optimum performance. The fabricated antenna is

tested and measured performance is compared with simulation.

This thesis includes 4 chapters. Chapter 2 consists of design and simulation of the

feed network, radiating element of the whole antenna. In Chapter 3, analysis of the

simulation results and measurements results are given. The reflection coefficient, gain,

radiation patterns and axial ratio are measured and compared with simulation. Finally in

Chapter 4, the discussions and conclusions are presented.

-3-

II. Antenna Design

2.1 Design Requirements

Portable data terminals are typically configured as hand-held computers, personal data

assistants(PDAs), laptop and notebook computers, and ruggedized data collectors.

Portable terminals have been traditionally used for collecting data via keyboard input bar

code scanner, magnetic strip reader and WiFi network connections. A typical UHF

portable RFID reader is shown in Fig.2.1.

Fig. 2.1 A typical portable RFID reader and its antenna

A suitable antenna for the portable terminals should be of low cost, low profile,

light weight and of especially small size. Antenna should have circularly-polarized

radiation. For the present study, the antenna design goal is listed in Table 2.1.

Table 2.1 Design goal of the proposed RFID reader antenna

Characteristics Requirements

Center frequency 912MHz

Bandwidth > 10MHz

Input VSWR < 3

Axial ratio < 3dB

Gain > 3dBic

Size < 62x62x15mm

Weight < 30g

-4-

2.2 Literature Review

In this section, a review of literature on light-weight and compact circularly-polarized

antennas is presented. Among many seemingly useful papers, one finds a rather limited

number of results applicable to the goal of this thesis. The size reduction is the single

most difficult requirement to meet.

Son and Lim studied a printed square quadrifilar spiral (QSA) antenna for use in

UHF RFID applications [1]. They used the fold inverted-F antenna in a spiral

configuration. Figure 2.2 shows their antenna geometry.

Fig. 2.2 Geometry of a compact printed square QSA [1]

The antenna is mounted on a grounded FR4 substrate, whose thickness is 0.7mm and

size is 60x60mm2 (Wg=60mm). At the reverse side of the substrate, the feed network is

implemented to produce four equal amplitudes and four different phases of 0°, 90°,

180° and 270°. Four inverted-F spirals of QSA are printed on a FR4 substrate and are

wound into a square shape. The antenna width (Wa) is 35mm and the height (H) is

18mm. The length of the spiral (Wc) is chosen to be 62mm, thickness of the spiral is

2mm. The distance between spiral arm and matching stub (Lm), the height of the

matching stub (Hm) are chosen to be 3mm and 7mm respectively. The authors also

presented an 2 by 2 array of QSA for use in fixed RFID reader applications.

Fig. 2.3(a) shows the reflection coefficient of a single QSA and an 2x2 array of

-5-

QSA. For a single QSA, the center frequency is 810MHz and the bandwidth of -10dB

reflection is 200MHz. Fig. 2.3(b) shows the measured peak gain. The single QSA has a

peak gain of 1.5dBic at 910MHz. Fig. 2.3(c) shows the measured axial ratio. The single

QSA has an axial raito of 2.75dB at 910MHz. Fig. 2.4 shows measured radiation

patterns of the single QSA. The QSA has broad beam widths and reduced radiation in

the backward directions.

(a)

(b) (c)

Fig. 2.3 Measured performance of QSA [1]. (a) Return loss, (b) gain,

and (c) axial ratio

-6-

Fig. 2.4 Measured radiation pattern of the single QSA at 910MHz [1]

Sung studied a circularly polarized square ring slot antenna with arrow-shaped

structure [2] shown in Fig. 2.5. The antenna is designed for operation at 1.4GHz and

can be scaled for 912MHz operation. White regions represent slots etched on ground

plane and the dotted line denote a microstrip feed line on the opposite side. To

maintain symmetry of the antenna, four arrows on the top, bottom, left, and right are

designed with the same shape shown in Fig 2.5(a). The antenna fabricated on an FR-4

substrate with h = 1.6mm, εr = 4.4 and tanδ = 0.01.

(a) (b)

Fig. 2.5 Arrow-shaped ring slot antenna for (a) linear and (b) circular polarizations [2]

-7-

(a) (b)

Fig. 2.6 Current distribution on the arrow-shaped ring slot antenna. (a) Linear

polarization and (b) circular polarization [2]

The antenna is fed by proximity coupling. The CP radiation characteristics are

achieved by loading with a proper asymmetry, which can be placed diagonally shown in

Fig 2.5(b). The difference between slot widths on different diagonal lines converts the

linearly-polarized radiation into a circularly-polarized one. The slot is of square shape

with the slot length L is 20mm. The ground plane size is either 40x40mm or 80x80mm

with inferior performance for smaller ground plane. Fig. 2.6 shows the current

distribution on the antenna surface at 1.38GHz where one can observe a clear difference

between linear and circular polarized operations.

Fig. 2.7 shows the reflection coefficient, radiation pattern and axial ratio of the

antenna. The overall size of the ground plane is 80×80mm. The center frequency is

1.38GHz, the -10dB reflection coefficient bandwidth is 2% and the 3dB axial ratio

bandwidth is 0.9%. At 1.38GHz, the antenna has a gain of 1.6dBic and an axial ratio

of 1.7dB. The ground plane size can be reduced from 80x80mm to 40x40mm with

resultant inferior performance. The 40x40mm ground plane size at 1.38GHz is scaled to

61x61mm at 912MHz.

-8-

(a)

(b) (c)

Fig. 2.7 Simulated and measured (a) reflection coefficient, (b) radiation pattern,

and (c) axial ratio of the arrow-shaped ring slot antenn [2]

2.3 Proposed Antenna Structure

The objective of this work is to design a small and lightweight antenna for portable

RFID reader applications. The required frequency range 907-917MHz is allocated for

UHF-band RFID applications in Korea. The design requirements are given in Table 2.1

of Section 2.1.

Among many possible structures, the one shown in Fig. 2.8(a) is finally chosen. The

antenna consists of four radiating elements shown in Fig. 2.8(c) and the feed network

of Fig. 2.8(d). The radiating element is connected to the feed network using the tuning

element shown in Fig. 2.8(b). The length of the antenna arm is reduced using

-9-

meander-line technique. The radiating element combined with the tuning element is

conceptually similar to a standard inverted-F antenna. Although the radiation pattern is

not fully uni-directional with a limited size of the ground plane, this structure offers a

higher gain than bi-direcional structures that radiate an equal amount of power in both

directions.

Radiating element

Gro

undi

ng

Tunning element

Feed network

Via hole

Shorting pin

Tuning element

(a) (b)

Radiating element

Printed meander monopole

Meander line quadrature coupler Input port

Isolated port Phas

e de

lay

line

Pow

er d

ivid

er

Ground plane

Output ports

Output ports

(c) (d)

Fig. 2.8 Proposed antenna structure

The feed network consists of a quadrature hybrid coupler and two power dividers. In

order to reduce the circuit size, quarter-wave lines in the quadrature coupler are

meandered and a dielectric cover is placed on top of the circuit. Successive 90° phase

shifts are obtained using the 90° phase difference in output arms of the coupler and by

properly adjusting lengths of the power divider output lines.

-10-

2.4 Feed network

The feed network consists of a quadrature coupler and two-way power dividers.

Theory in Pozar's book states that quadrature couplers are 3dB directional couplers with

a 90° phase difference in the outputs of through and coupled arms [3]. The quadrature

coupler shown in Fig. 2.9 is often made in microstrip or stripline form.

0Z

0Z

0Z

0Z

0Z0Z

0 2Z

4

l

4

l

Input (port 1)

Isolated (port 4)

Output (port 2)

Output (port 3)

0 2Z

Fig. 2.9 Geometry of a quadrature coupler [3]

The basic operation of the branch-line quadrature coupler is as follows. With all

ports matched, power entering port 1 is evenly divided between port 2 and 3, with a

90° phase difference between these outputs. No power is coupled port 4 (the isolated

port). Thus, the scattering matrix of an ideal quadrature coupler will have the following

form:

(2.1)

Observe that the quadrature coupler has a high degree of symmetry, as any port can be

used as the input port [3].

First a standard un-miniaturized quadrature coupler shown in Fig. 2.10 is investigated

using a commercial electromagnetic simulation software Microwave StudioTM (MWS) by

CST. The quadrature coupler has four circuit layers placed on two printed circuit boards

(PCB's), which are bonded together. The upper board is employed to increase the

effective dielectric constant of the microstrip line. The thickness of each board is 0.5mm

-11-

and of FR-4 type with εr = 4.6 and tanδ = 0.02 at 900MHz frequency band. Board 1

has a bottom layer which is a ground plane for the radiating element as well as for the

hybrid coupler, and a top layer which has the quadrature coupler circuit pattern. Board

2 is just a cover and has no metal layers. The bare PCB has 1-oz. copper metallization

so that the metal thickness is 0.034mm.

2V

3V

1V

4V

4

l

4

l

2w

1w

50 mm

50

mm

1re

2reQuadrature coupler

circuit

Ground

2h

1hBoard 1Board 2

Fig. 2.10 A simple quadrature coupler structure

The characteristic impedances of dielectric-covered microstrip lines are obtained using

MWS. Figure 2.11 shows scattering parameters of the simple quadrature coupler. The

reflection coefficient is less than -20dB over 880-930MHz (5.5% bandwidth) and |S11| =

-22 dB at 912MHz. Transmission coefficients |S31| and |S21| are both -3.5dB at 912

MHz. The additional 0.5dB loss is due to the line loss in the microstrip line on the

FR-4 substrate. The transmission coefficient from port 1 to the isolated port 4 is less

than -20dB at 860-970MHz and is -41dB at 912MHz.

-12-

800 850 900 950 1000-50

-40

-30

-20

-10

0

Mag

nitu

de

(dB

)

Frequency (MHz)

|S11

|

|S41

|

|S31

|

|S21

|

Fig. 2.11 Scattering parameters of a simple quadrature coupler

800 850 900 950 1000-210

-180

-150

-120

-90

-60

Ph

ase

(deg

)

Frequency (MHz)

arg(S21

)

arg(S31

)

Fig. 2.12 Phase of the transmission coefficient of a simple quadrature coupler

Figure 2.12 shows the phase of the transmission coefficient of a simple quadrature

coupler. Insertion phases arg(S21) and arg(S31) are -92° and -182° at 912 MHz,

respectively, so that the phase difference arg(S31)-arg(S21) is -90° as shown in Fig 2.13.

-13-

800 850 900 950 1000-120

-105

-90

-75

-60

Pha

se (

deg)

Frequency (MHz)

arg(S31

)-arg(S21

)

Fig. 2.13 Phase difference of a simple quadrature coupler

The simple quadrature coupler shown in Fig. 2.10 occupies an area of 50x50mm2. In

order to feed four radiating elements, one has to include power dividers and delay lines

which cannot be realized in an area left over by the quadrature coupler out of the

desired area 62x62mm2. Therefore the size of the quadrature coupler should be reduced.

One way to this is using meandered lines as shown in Fig. 2.14.

Quadrature coupler circuit

2V

1V

4V

3V

1w

2w

4

l

4

l

4

l

4

l

Fig. 2.14 Miniaturized quadrature coupler

-14-

Fig. 2.15(a) shows the cross section of layered substrates of the miniaturized coupler.

The bottom layer of board 1 has a ground plane while the top layer has no

metalization. The quadrature coupler is placed on the bottom layer of board 2.

Grounding strips are placed on the top layer of board 2 along the rectangular edge. Fig.

2.15(b) shows two boards bonded together.

Board 1

Board 2Top layout

Bottom layout

Top layout

Bottom layout

2re

2h

1h1re

L


Ground

(a)

BoundedBoard2

+Board1

h


(b)

Fig. 2.15 Cross section of the miniaturized quadrature coupler circuit. (a) Board

construction and (b) two boards bonded together

Line widths, characteristic impedances, and effective dielectric constants of the

dielectric covered microstrip lines used in the miniaturized quadrature coupler are listed

in Table 2.2.

Table 2.2 Parameters of microstrip lines used in the miniaturized quadrature coupler

LinesLine widths

(mm)

Characteristic

impedance (Ohms)

Effective dielectric

constant (εr)

w1 0.74 50 3.72w2 1.42 35 3.66

w3 0.14 100 3.89

-15-

In order to design the miniaturized quadrature coupler shown in Fig. 2.10, one has

to find high-performance meander lines. Fig. 2.16 shows a 50-ohm bended line of

quarter-wavelength used in branch lines of the miniaturized quadrature. Circular

chamfering is applied at right-angle bends in order to minimize the reflection due to

discontinuities presented by the bends.

The reflection and transmission coefficients of the 50-ohm bended line are shown

in Fig. 2.17 and 2.18. At 912MHz, the reflection coefficient is -33dB and the

transmission coefficient is -0.35dB. The phase of the transmission coefficient is -90°. A

loss of 0.35dB is attributed to the loss tangent of 0.02 of the FR-4 substrate.

Fig. 2.16 50-ohm bended line of quarter-wavelength

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de (

dB)

Frequency (MHz)

|S11

|

Fig. 2.17 Reflection coefficient of 50-ohm bended line

-16-

800 850 900 950 1000-2.0

-1.5

-1.0

-0.5

0.0

Mag

nitu

de (

dB

)

Frequency (MHz)

|S21|

800 850 900 950 1000-120

-105

-90

-75

-60

Pha

se (

deg

)

Frequency (MHz)

arg(S21)

(a) (b)

Fig. 2.18 Transmission coefficient of 50-ohm bended line. (a) Magnitude and (b) phase

Fig. 2.19 35-ohm bended line of quarter-wavelength

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nit

ude

(d

B)

Frequency (MHz)

|S11

|

Fig. 2.20 Reflection coefficient of 35-ohm bended line

Fig. 2.19 shows a 35-ohm bended line of quarter-wavelength used in main lines of

the miniaturized quadrature coupler. Its reflection and transmission coefficients are shown

in Fig. 2.20 and 2.21. At 912MHz, the reflection coefficient is -34dB and the

-17-

transmission coefficient is -0.34dB. The phase of the transmission coefficient is -90°. A

loss of 0.34dB is attributed to the loss tangent of 0.02 of the FR-4 substrate.

800 850 900 950 1000-2.0

-1.5

-1.0

-0.5

0.0

Mag

nit

ud

e (d

B)

Frequency (MHz)

|S21

|

800 850 900 950 1000-120

-105

-90

-75

-60

Pha

se (

deg)

Frequency (MHz)

arg(S21)

(a) (b)

Fig. 2.21 Transmission coefficient of 35-ohm bended line. (a) Magnitude and (b) phase

Combining the bended lines of Figs. 2.16 and 2.19, the miniaturized quadrature

coupler shown in Fig. 2.14 is designed. Fig. 2.22 shows the reflection coefficient of the

miniaturized quadrature coupler. The reflection coefficient of port 1 is less than -20dB

over 860-950 MHz (bandwidth 9.8%) and is -27 dB at 912MHz while that of port 2 is

less than -20dB over 880-970 MHz (bandwidth 9.8%) and is -26 dB at 912MHz.

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de (

dB)

Frequency (MHz)

|S11

|

|S22

|

Fig. 2.22 Reflection coefficient of the miniaturized quadrature coupler

-18-

800 850 900 950 1000-60

-50

-40

-30

-20

-10

0

Mag

nitu

de

(dB

)

Frequency (MHz)

|S41

|

|S31

|

|S21

|

Fig. 2.23 Transmission coefficient of the miniaturized quadrature coupler

Fig. 2.23 shows the transmission coefficient of the miniaturized quadrature coupler.

The transmission coefficient of port3 and port2 is approximately -3.6dB at 912MHz.

The transmission coefficient of between port 1 and isolated port 4 is less than -20dB

over 860-970MHz and is -30dB at 912MHz. Fig. 2.24 shows the phase of the

transmission coefficient where arg(S21)=-94° and arg(S31)=-184° at 912MHz. Phase

difference arg(S31)-arg(S21) is -90° at 912MHz as shown in Fig. 2.25.

800 850 900 950 1000-210

-180

-150

-120

-90

-60

Ph

ase

(deg

)

Frequency (MHz)

arg (S21)

arg (S31)

Fig. 2.24 Transmission phase of the miniaturized quadrature coupler

-19-

800 850 900 950 1000-120

-105

-90

-75

-60

Pha

se (

deg)

Frequency (MHz)

arg(S31

)-arg(S21

)

Fig. 2.25 Transmission phase difference of the miniaturized quadrature coupler

The feed network based on the miniaturized quadrature coupler is shown in Fig.

2.26. At ports 2 and 3 of the coupler, a two-way power divider is implemented so that

four 100-ohm lines are obtained for feeding four radiating elements. The length of the

lines for V5 and V6 are one guided wavelength so that a proper set of phasing is

obtained for circular polarization. Assuming the phase of V1 is zero degree, nominal

phases of V2, V3, V6, and V5 are -90°, -180°, -270°, -360° which are proper values for

circularly-polarized radiation. V4 is loaded with a 50-ohm resistor.

V5V2

V3

V1

V6

V4

(a)

-20-

B o ard 1

B o ard 2T o p lay o u t

B o tto m lay o ut

T op lay o u t

B otto m lay o u t

2re

2h

1h1re

L

F ee d n etw o rk c ircu it

G ro un d1V4V

3V 6V

via h ole

5V2V

(b)

Board2+

Board1h

1V

5V3V

4V

Output (port 2,3) Output (port 5,6)

Izolated (port 4) Input (port 1)via hole

Feed network circuit

2V 6V

(c)

Fig. 2.26 Antenna feed network. (a) Circuit layout, (b) two boards unbonded, and (c)

two boards bonded together

Input port V1 and isolated port V4 are routed on to the bottom layer of the lower

board using aligned via holes in both board 1 and board 2. Aligned via holes are

utilized to ensure a proper electrical connection from the quadrature coupler on the

bottom layer of board 2 and the port terminal on the bottom layer of board 1. Output

ports V2, V3, V5, and V6 are routed to the top surface of board 2. Along the edge of the

top surface of the board 2, a strip of ground plane is placed and electrically connected

to the ground plane on the bottom face of board 1 with closely-spaced via holes.

Fig. 2.27 shows the reflection coefficient of the feed network. The reflection

coefficient is -20dB over 880-930MHz and is -23dB at 912MHz. Fig. 2.28 shows the

transmission coefficient of the feed network. At 912MHz, transmission coefficients from

port 1 to ports 2, 3, 5 and 6 are -5.5dB, -5.8dB, -7.7dB, -7.4dB, respectively.

Assuming that the line loss per a quarter wavelength is 0.35dB, nominal values of

-21-

transmission coefficients from port 1 to ports 2, 3, 5, and 6 are -6.35dB, -6.70dB,

-7.40dB, and -7.05dB, respectively. Deviation from the nominal value is attributed to the

imperfect operation of the power divider.

800 850 900 950 1000-25

-20

-15

-10

-5

0

Mag

nit

ude

(dB

)

Frequency (MHz)

|S11

|

Fig. 2.27 Reflection coefficient of the feed network

800 850 900 950 1000-10.0

-7.5

-5.0

-2.5

0.0

Mag

nitu

de (

dB

)

Frequency (MHz)

|S21

|

|S31

|

|S51

|

|S61

|

Fig. 2.28 Transmission coefficient of the feed network

Fig. 2.29 shows the transmission phase of the feed network, where arg(S21)=-90.0°,

arg(S31)=-177.7°, arg(S51)=-359.8°, and arg(S61)=-269.0° at 912MHz. Transmission phases

-22-

are de-embedded so that a phase delay due to common input and output lines, about

15°, is removed. Fig. 2.30 shows the transmission phase deviation of S21, S31, S61, and

S51 from nominal values of -90°, -180°, -270°, and -360°, respectively. One can see that

phase deviation is ±2.5° at 912MHz.

800 850 900 950 1000

-360

-270

-180

-90

0

arg(S61

)=-269o @ 912 MHz

arg(S51

)=-359.8o @ 912 MHz

arg(S31

)=-177.7o @ 912 MHz

arg(S21

)=-90o @ 912MHz

Ph

ase

(deg

)

Frequency (MHz)

Fig. 2.29 Transmission phase of the feed network

900 905 910 915 920 925-10

-5

0

5

10

Ph

ase

(deg

)

Frequency (MHz)

arg(S21

)

arg(S31

)

arg(S51

)

arg(S61

)

Fig. 2.30 Transmission phase deviation of the feed network

-23-

2.5 Radiating elements

(1) Design of a single element

The antenna element employed in this thesis is a meander-line inverted F antenna

shown in Fig. 2.31 which has a small size and a good impedance property. The

meander line makes it possible to significantly reduce the size of the radiating element

while the tuning stub improves the impedance matching. The meander-line inverted F

antenna can be placed along the edges of the feed network circuit board, In this case,

however, one can expect a reduced gain, since the antenna radiates equally in both

directions.

l

Fig. 2.31 Single radiating element

In designing a single element, the feed network is not used but a discrete excitation

with a source impedance of 100Ω is employed between the feed arm terminal and the

ground plane. The tuning stub is short-circuited.

Fig. 2.32 shows the dependence of the reflection coefficient on the length of the

meander line l where one can see that the resonant frequency can be tuned by varying

the meander line length. With l=0.323λ, a resonance is obtained at the 912MHz.

Fig. 2.33 shows the reflection coefficient of a single element with or without a shorting

pin. In this case, the length of meander line is l=0.282λ. One can observe in Fig. 2.33

that there is a dramatic effect of the shorting stub on the impedance matching.

-24-

800 900 1000 1100 1200-20

-15

-10

-5

0

Mag

nit

ude

(d

B)

Frequency (MHz)

l=0.282l l=0.301l

l=0.323l

Fig. 2.32 Reflection coefficient of a single element versus antenna length

1000 2000 3000-20

-15

-10

-5

0

Mag

nit

ud

e (d

B)

Frequency (MHz)

|S11

| (l=0.282l, single element with shorting pin )

|S11

| (I=0.282l, single element without shorting pin)

Fig. 2.33 Impedance matching of a single element by the shorting stub

Fig. 2.34 shows the three-dimensional gain patterns of a single element antenna. Fig.

2.34(a) is an absolute gain. The maximum value of the absolute gain is -1.8dBi and the

radiation efficient is 39%. Figs. 2.34(b) and (c) show the gains of the theta and phi

components, respectively.

Fig. 2.35 shows two-dimensional total gain patterns of a single element. The total

gain is of an omni-directional shape on the zx-plane and of a nearly omni-directional

shape on xy- and yz-plane.

-25-

(a)

(b)

(c)

Fig. 2.34 3D gain patterns of a single radiating element. (a) Total gain, (b) gain of

theta component, and (c) gain of phi component

-26-

(a)

(b)

(c)

Fig. 2.35 2D total gain patterns of a single radiating element. (a) On zx-plane, (b) on

xy-plane, and (c)on yz-plane

-27-

Next the performance of a four-element array shown in Fig. 2.36 is investigated using

a idealized feeding. Each element is fed with a voltage of the same magnitude and of a

successive 90-degree phase shift. The source impedance of each port is 100 ohms.

V3

V2

V5

V6

Fig. 2.36 Array of four radiating elements

When a four-element array is formed, there is a significant level of mutual coupling.

Fig. 2.37 shows the change in the reflection coefficient of a single element when there

are three other elements terminated in 100Ω. The resonance frequency is 1020MHz

when there is a single element alone, while it is 912MHz when three other elements

are added. Thus it is evident that one has to tune the single element in the presence of

three other elements. In fact in order to obtain Fig. 2.37, the length of the radiating

element is adjusted so that it resonates with all other elements present.

Fig. 2.38 shows the mutual coupling between elements. The mutual coupling between

two parallel elements such as V2 and V6 is about -5dB, while it is about -12.5dB

between orthogonal elements such as V2 and V5. The level of the mutual coupling is

fairly high due to the close proximity between elements.

Fig. 2.39 shows the 3D radiation pattern of the four-element array with an excitation

ideal for circular polarization, i.e., with voltages of the same magnitude and of

successive 90-degree phase difference. At 912MHz the total gain is 3.25dBic, and the

radiation efficient is 0.78%. The gain of theta and phi components is 0.25dBi. Theta

-28-

and phi components of the radiation are combined in phase quadrature (i.e., with a

90-degree phase difference) to form a circularly polarized radiation with a gain of

3.25dBic.

800 900 1000 1100 1200-20

-15

-10

-5

0

Mag

nit

ude

(dB

)

Frequency (MHz)

|S11

| (l=0.282l, single element with other 3 elements)

|S11

| (l=0.282l, single element alone)

Fig. 2.37 Change in the resonance frequency of a single element due to mutual

coupling

800 850 900 950 1000-30

-25

-20

-15

-10

-5

0

Mag

nitu

de (

dB)

Frequency (MHz)

|S22

|

|S32

|

|S52

|

|S62

|

Fig. 2.38 Mutual coupling between antenna elements

-29-

(a)

(b)

(c)

Fig. 2.39 3D gain pattern of the four-element array. (a) Total gain, (b) gain of theta

component, and (c) gain of phi component

-30-

(a)

(b)

(c)

Fig. 2.40 2D total gain patterns of the four-element array. (a) On zx-plane, (b) on

yz-plane, and (c) on xy-plane

-31-

Fig. 2.40 shows the 2D total gain patterns of the four-element array. The antenna

has about a 7dB front-to-back ratio, which is not high due to the small ground plane

size. In the horizontal direction, the array has an omni-directional pattern. The antenna

has a 126-degree 3-dB beamwidth.

Fig. 2.41 shows the 3D axial ratio and polarization patterns of the four-element

array. In +z axis (the antenna boresight) the antenna has an axial ratio less than 1dB.

Figs. 2.41(a) and (b) show the right- and left-hand circular polarization patterns. Due to

a good axial-ratio performance, the cross polarization discrimination of the antenna is

excellent. Fig. 2.42 shows the normalized axial pattern of the four-element array. The

3-dB axial ratio beamwidth is about 120 degrees on both planes.

-32-

(a)

(b)

(

(c)

Fig. 2.41 3D polarization pattern of the four-element array. (a) Axial ratio, (b) right-

hand circular polarization, and (c) left-hand circular polarization

-33-

(a)

(b)

Fig. 2.42 2D axial ratio pattern. (a) On zx-plane and (b) on yz-plane

-34-

2.6 Whole Antenna Structure

The final four-element antenna is obtained by combing the four radiating elements

with the feed network as shown in Fig. 2.43, where the upper circuit board is rendered

in a transparent form to enhance the readability. The meander line length of the

radiating element is l=0.282λ.

Fig. 2.43 Final four-element antenna structure

Fig. 2.44 shows the reflection coefficient of the final antenna structure. The

reflection coefficient of the antenna is less than -10 dB over 810-1000MHz and is -16

dB at 912MHz. Fig. 2.45 shows the gain versus frequency. The gain at 912MHz is

2.4dBic, which is 0.85dB less than that of the antenna fed by idealized excitation. The

0.85dB loss is mostly due to the transmission line loss arising from a dielectric loss in

the FR-4 substrate. Fig. 2.46 shows the axial ratio versus frequency. The axial is less

than 0.6dB over 900-920MHz. The wide-band performance of the axial ratio is due to

the inherent property of the feed network.

Fig. 2.47 shows the 3D gain pattern of the final antenna. The maximum gain is

2.4dBic and the radiation efficiency is 64%. Gain of theta and phi components is -0.04

dB. Fig. 2.48 shows the 2D total gain pattern of the final antenna. The front-to-back

ratio is about 7dB. The antenna has an omni-directional pattern in the horizontal plane.

-35-

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de (

dB

)

Frequency (MHz)

|S11|

Fig. 2.44 Reflection coefficient of the final antenna

900 905 910 915 9200

1

2

3

Gai

n (

dB)

Frequency (MHz)

Simulation

Fig. 2.45 Gain versus frequency of the final antenna

900 905 910 915 9200

1

2

3

Ax

ial

rati

o (

dB)

Frequency (MHz)

Simulation

Fig. 2.46 Axial ratio versus frequency of the final antenna

-36-

(a)

(b)

(b)

Fig. 2.47 3D gain pattern of the final antenna. (a) Total gain, (b) gain of theta

component, and (c) gain of phi component

-37-

(a)

(b)

(c)

Fig. 2.48 2D total gain pattern of the final antenna. (a) On zx-plane,

(b) on yz-plane, and (c) on xy-plane

-38-

(a)

(b)

(c)

Fig. 2.49 3D polarization pattern of the final antenna. (a) Axial ratio, (b) right-hand

circular polarization, and (c) left-hand circular polarization

-39-

(a)

(b)

Fig. 2.50 2D axial ratio pattern of the final antenna. (a) On zx-plane and (b) on yz-

plane

Fig. 2.49 shows the 3D axial ratio and polarization pattern of the final antenna. The

axial ratio is less than 1dB in +z direction. The whole antenna has a good axial ratio

performance. Fig. 2.50 shows the 2D axial ratio pattern. The 3dB axial ratio

beamwidths are about 100° in one plane and 140° in the other plane.

-40-

III. Antenna Fabrication and Measurements

3.1 Antenna Fabrication on PCB

Photo etching process is applied to fabricate the designed antenna on FR-4 substrate.

Fig. 3.1 shows the fabricated antenna components before assembly. The dimension of

the components is as follows. Feed network board: 60.0x60.0x1.0mm, radiating elements

board: 50.0x50.0x0.5mm, and tuning element board: 13.0x8.5x0.5mm.

(a) (b)

(c)

Fig. 3.1 Photographs of the fabricated antenna components. (a) Feed network,

(b) radiating elements, and (c) tuning elements

Fig. 3.2 shows the photograph of the fabricated and assembled antenna. The upper

circuit board contains four radiating elements while the lower one has the feed network.

-41-

The two boards are fastened by the vertically-installed tuning elements and by soldering.

Fig. 3.2 Photograph of the fabricated antenna

3.2 Antenna Tuning

Antenna is tuned for better performance after fabrication. Tuning improves the antenna

impedance matching and gain. Although one can expect that the design using MWS is

accurate enough to eliminate the need for antenna tuning, the practice is different. Apart

from finite accuracy of the numerical modelling by MWS, there are certain factors that

cannot completely be accounted for, among which are inaccurate dielectric constant of

the FR-4 substrate and finite conductivity of copper cladding.

First the length of the meander line radiating element is adjusted. The lengths of all

radiating elements is changed by the same amount d as shown in Fig. 3.3. Reducing

the element length by 2.5mm gives a good result. Fig. 3.4 shows the improvement in

the reflection coefficient by adjusting the element length. Increasing the element length

also improves the reflection coefficient but the gain is reduced.

-42-

ld

d

d

d

Fig. 3.3 Antenna tuning by adjusting the length of radiating elements

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de (

dB)

Frequency (MHz)

l= 0.290l

l= 0.282l

Fig. 3.4 Improvement in reflection coefficient by adjusting the length of radiating

elements

Next the strip width of the tuning circuit is adjusted for better performance. Fig. 3.5

shows the improvement in the impedance matching when the width of strip in the

tuning circuit is increased from 0.25mm to 0.8mm. Also about 1dB increase in gain is

obtained by adjusting the tuning circuit strip width.

-43-

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de (

dB)

Frequency (MHz)

w4=0.25mm

w4=0.8mm

Fig. 3.5 Impedance tuning by modifying tuning strip width

3.3 Antenna Measurement

After the antenna tuning is finished, the various antenna characteristics are measured.

Fig. 3.6 shows the measured reflection coefficient of the fabricated and tuned antenna.

800 850 900 950 1000-40

-30

-20

-10

0

Mag

nitu

de

(dB

)

Frequency (MHz)

Simulation Measurement

Fig. 3.6 Reflection coefficient of the fabricated antenna

-44-

The measured reflection coefficient is less than -10dB over 880-1000MHz (13.1%

bandwidth) and is -16dB at 912MHz. There is a significant discrepancy between

measured and simulated reflection coefficients.

Fig. 3.7 shows the measured gain versus frequency. At 912MHz, the antenna gain is

3.2dBic. There is a considerable amount of difference between measured and simulated

gains.

900 905 910 915 9200

1

2

3

4

5

Gai

n (d

Bic

)

Frequency (MHz)


Fig. 3.7 Gain versus frequency of the fabricated antenna

Fig. 3.8 shows the comparison of the gain of the fabricated antenna with those of

existing antennas shown in Fig. 3.9. The fabricated antenna has a gain value similar to

that of an antenna by Actenna Co. The ceramic patch antenna by MAC technologies

has a gain of 4.3dBic, about 1dB greater than the fabricated antenna.

Fig. 3.10 shows the gain pattern of the fabricated antenna. The measured gain pattern

agrees well with simulation. Fig. 3.11 shows the axial ratio versus frequency of the

fabricated antenna. The measured axial ratio is less than 1dB over 906-920MHz and is

0.6dB at 912MHz. There is some difference in measured and simulated axial ratios. The

performance of the fabricated antenna is summarized in Table 3.1.

-45-

900 905 910 915 9200

1

2

3

4

5

G

ain

(d

Bic

)

Frequency (MHz)

Our fabricated antenna Actenna Co. MAC Technologies

Fig. 3.8 Comparison of gain of the fabricated antenna with those of existing antennas

(a) (b)

Fig. 3.9 Photographs of existing RFID reader antenna. (a) One by Actenna Co.(size:

60x60x15mm), and (b) ceramic patch antenna by MAC Technologies (size:

62x62x7mm)

-46-

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150180

210

240

270

300

330

-30

-25

-20

-15

-10

-5

0

5


(a)

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150180

210

240

270

300

330

-30

-25

-20

-15

-10

-5

0

5


(b)

-30

-25

-20

-15

-10

-5

0

50

30

60

90

120

150180

210

240

270

300

330

-30

-25

-20

-15

-10

-5

0

5


(c)

Fig. 3.10 Measured gain pattern of the fabricated antenna. (a) On zx-plane,

(b) on yz-plane, and (c) on xy-plane

-47-

900 905 910 915 9200

1

2

3

Axi

al r

atio

(dB

)

Frequency (MHz)


Fig. 3.11 Axial ratio versus frequency of the fabricated antenna

Table 3.1 Performance of the fabricated antenna

Performance items Performance

Gain 3.2dBicOperating frequency 908-916MHz

Reflection coefficient -16dBAxial ratio 0.6dB

3dB beamwidth 100°/120°

Front-to-back ratio 7dBWeight 15.3g

Size 60x60x14mm

-48-

IV. Conclusions

In this thesis, a light-weight and compact circularly polarized antenna is developed

for the portable UHF RFID reader in order to replace the existing ceramic patch

antenna weighing over 70 grams. The antenna consists of an array of inverted-F type

meander line printed monopoles fed in successive 90° phase shifts by a feed network

where a quadrature hybrid coupler and two power dividers are employed. The feed

network placed on the lower circuit board while the radiating elements are located on

the upper board. The feed network and the radiating elements are connected via vertical

feed lines with impedance-matching shorting stub.

Sizes of the radiating element and the quadrature hybrid coupler are reduced using

the meander-line technique. The antenna is designed using the commercial software

Microwave StudioTM (MWS) by CST.

The designed antenna is fabricated by photo-etching antenna circuit patterns on the

common FR-4 PCB board. The length of the radiating elements and width of the

vertical feed lines are adjusted for optimum performance at 912MHz. Measurements

show that the fabricated antenna has an axial ratio of 0.6dB, a gain of 3.2dBic, and a

reflection coefficient less than -16dB at 912MHz. The antenna size is 60x60x14mm and

the antenna weight is 15.3g. The performance of the developed antenna is good enough

to replace the existing heavier patch antenna for the portable UHF RFID applications.

The antenna concept presented in this can also be applied for other applications where

a small light-weight circularly-polarized antenna is required.

-49-

References

[1] W. I. Son, W. G. Lim, M. Q. Lee, S. B. Min, and J. W. Yu "Printed square

quadrifilar antenna for UHF RFID reader," IEEE AP-S Int. Symp. Digest, 9-15 June

2007, pp. 305-308.

[2] Y. Sung, "Circularly polarized ring slot antenna with arrow-shaped structure," ETRI

Jour., pp. 506-509, Vol. 31, No. 5, Oct. 2009.

[3] D. M. Pozar, Microwave Engineering, 2nd Ed., New York: John Wiley & Sons,

1998.

-50-

Acknowledgements

For three years of learning a lot of things and studying in Applied Electromagnetic

Laboratory at the Chungbuk National University in South Korea, I would like to give

special thanks to my advisor Professor Bierng-Chearl Ahn. He helped me a lot during

my study. I studied and knew many fundamental theories and practical knowledge in

RF engineering and antenna design.

Then, I wish to thank Professor Jae-Hyung Ahn, the chairman of my thesis review

committee, and Professor Kyung-Seok Kim, a member of my thesis review committee.

Both Professors gave me valuable advice for improving my thesis. I feel that this thesis

is the most important part of my study in Master's program. Actually I will be

graduating from the Department of Bio and Information Technology with a Master's

Degree in Bio-electronics. However I studied very much in radio engineering field from

Laboratory students and from Research Professor Jae-Hoon Bang.

Special thanks should go to Khongorzul Dashdondov for starting me up with

everything. I thank my friend Battseren Sharavsambuu, who has been living with me for

three years. Spending difficult time together is unforgettable. I also thank Bayanmunkh

Enkhbayar and Ononchimeg Sodnomtseren who are Ph.D. students in the Laboratory.

They gave me a lot of help on the design and simulation of the antenna.

I would like to thank BIT Research-Oriented University Consortium of Chungbuk

National University for financial support during my graduate study.

Finally, I would like to mention my family: Father Bat-Ochir, Mother Buyankhishig,

oldest brother Avirmed, older brother Enkhbaatar, sister Badamkhand and a younger

brother Batkhishig. My wonderful family, I love all of you.

2009-12-20 in Lab.

Chinzorig Bat-Ochir

I. IntroductionII. Antenna Design 2.1 Design Requirements 2.2 Literature Review2.3 Proposed Antenna Structure 2.4 Feed Network 2.5 Radiating Elements 2.6 Whole Antenna Structure

III. Antenna Fabrication and Measurements3.1 Antenna Fabrication on PCB3.2 Antenna Tuning3.3 Antenna Measurements

IV. ConclusionsReferences

11I. Introduction1II. Antenna Design 3 2.1 Design Requirements 3 2.2 Literature Review4 2.3 Proposed Antenna Structure 8 2.4 Feed Network 10 2.5 Radiating Elements 23 2.6 Whole Antenna Structure34III. Antenna Fabrication and Measurements40 3.1 Antenna Fabrication on PCB40 3.2 Antenna Tuning41 3.3 Antenna Measurements43IV. Conclusions48References49

工學碩士學位論文ael.chungbuk.ac.kr/ael-results/theses/... · 2015-07-07 ·...

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